Radar apparatus

ABSTRACT

A radar apparatus includes a plurality of transmit antennas that transmits a plurality of transmission signals using a multiplexing transmission, and a transmission circuit that applies phase rotation amounts corresponding to combinations of Doppler shift amounts and code sequences to the plurality of transmission signals. Each of the plurality of transmission signals is assigned a different combination among the combinations. The combinations include at least one combination of different numbers of multiplexing by the code sequences.

CROSS-REFERENCE TO RELATED APPLICATIONS

This is a continuation of U.S. patent application Ser. No. 16/898,856,filed Jun. 11, 2020, which claims the benefit of priority of JapanesePat. Appl. No. 2019-110522, filed Jun. 13, 2019, and of Japanese Pat.Appl. No. 2019-221249, filed Dec. 6, 2019. The entire disclosure of eachof the above-identified documents is incorporated herein by reference inits entirety.

TECHNICAL FIELD

The present disclosure relates to a radar apparatus.

BACKGROUND ART

Radar apparatuses that use radar transmission signals with shortwavelengths, including microwaves or millimeter-waves that realize highresolution, have been under study in recent years. To improve safety inoutdoor spaces, a demand has arisen for the development of radarapparatuses (referred to as, for example, wide-angle radar apparatuses)for sensing vehicles, as well as small objects such as pedestrians,within wide-angle ranges.

Examples of the configuration of a radar apparatus having a wide-anglesensing range include a configuration that uses a method calleddirection of arrival (DOA) estimation. In DOA estimation, reflectedwaves from a target are received by an array antenna constituted by aplurality of antennas (or also referred to as antenna elements), and thedirection of arrival (or referred to as the angle of arrival) of thereflected waves is estimated using a signal processing algorithm basedon reception phase difference as to element spacing (antenna spacing).Techniques for DOA estimation include, for example, the Fourier method,or examples of the technique for realizing high resolution include theCapon method, multiple signal classification (MUSIC), and estimation ofsignal parameters via rotational invariance techniques (ESPRIT).

Further, for example, a configuration of a radar apparatus (alsoreferred to sometimes as a multiple input multiple output (MIMO) radar)has been proposed that includes a plurality of antennas (array antenna)on the transmitting side in addition to the receiving side such that thetransmit and receive array antennas are used to perform signalprocessing to perform beam scanning (see, for example, NPL 1).

CITATION LIST Patent Literature PTL 1

-   Japanese Patent Application Laid-Open No. 2008-304417

PTL 2

-   Japanese Unexamined Patent Application Publication (Translation of    PCT Application) No. 2011-526371

PTL 3

-   Japanese Patent Application Laid-Open No. 2014-119344

Non Patent Literature NPL 1

-   J. Li, and P. Stoica, “MIMO Radar with Colocated Antennas”, Signal    Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007

NPL 2

-   M. Kronauge, H. Rohling, “Fast two-dimensional CFAR procedure”, IEEE    Trans. Aerosp. Electron. Syst., 2013, 49, (3), pp. 1817-1823

NPL 3

-   Direction-of-arrival estimation using signal subspace modeling    Cadzow, J. A.; Aerospace and Electronic Systems, IEEE Transactions    on Volume: 28, Issue: 1 Publication Year: 1992, Page(s): 64-79

NPL 4

-   V. Winkler, “Novel Waveform Generation Principle for Short-Range    FMCW-Radars,” in Proc. German Microw. Conf., 2009, pp. 1-4.

SUMMARY

However, methods for sensing a target object (or target) using a radarapparatus (for example, a MIMO radar) have not been fully studied.

A non-limiting exemplary embodiment of the present disclosure provides aradar apparatus with improved target-object sensing accuracy.

A radar apparatus according to one example of the present disclosureincludes: a plurality of transmit antennas that transmit a plurality oftransmission signals using a multiplexing transmission; and atransmission circuit that applies phase rotation amounts correspondingto combinations of Doppler shift amounts and code sequences to theplurality of transmission signals, wherein each of the combinations ofthe Doppler shift amounts and the code sequences has at least onedifferent from other combination, and wherein the number of multiplexesof the code sequence corresponding to at least one of the Doppler shiftamounts in the combinations is different from the number of multiplexingof code sequences corresponding to the remaining Doppler shift amount.

It should be noted that general or specific embodiments may beimplemented as a system, an apparatus, a method, an integrated circuit,a computer program or a storage medium, or any selective combination ofthe system, the apparatus, the method, the integrated circuit, thecomputer program, and the storage medium.

According to an exemplary embodiment of the present disclosure, thetarget-object sensing accuracy of a radar apparatus can be improved.

Additional benefits and advantages of the disclosed exemplaryembodiments will become apparent from the specification and drawings.The benefits and/or advantages may be individually obtained by thevarious embodiments and features of the specification and drawings,which need not all be provided in order to obtain one or more of suchbenefits and/or advantages.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram illustrating an example configuration of aradar apparatus according to Embodiment 1;

FIG. 2 is a diagram illustrating an example of transmission signals andreflected wave signals when chirp pulses are used;

FIGS. 3A and 3B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 4A and 4B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 5A to 5C are diagrams illustrating an example of the assignment ofDoppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 6A to 6C are diagrams illustrating an example of the assignment ofDoppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 7A and 7B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 8A and 8B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 9A and 9B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 10A and 10B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 11A and 11B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Embodiment 1;

FIGS. 12A and 12B are diagrams illustrating an example Doppler domaincompression process;

FIG. 13 is a diagram used to describe the Doppler domain compressionprocess;

FIGS. 14A and 14B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Embodiment 1;

FIG. 15 is a block diagram illustrating an example configuration of aradar transmitter according to Variation 3 of Embodiment 1;

FIG. 16 is a block diagram illustrating an example configuration of aradar apparatus according to Variation 5 of Embodiment 1;

FIGS. 17A and 17B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Variation 7of Embodiment 1;

FIGS. 18A and 18B are diagrams illustrating an example of the assignmentof Doppler shift amounts and orthogonal codes according to Variation 7of Embodiment 1;

FIG. 19 is a block diagram illustrating an example configuration of aradar apparatus according to Variation 8 of Embodiment 1;

FIG. 20 is a diagram illustrating an example of transmission signalswhen chirp pulses according to Variation 8 of Embodiment 1 are used;

FIG. 21 is a block diagram illustrating an example configuration of aradar apparatus according to Variation 9 of Embodiment 1;

FIG. 22 is a diagram illustrating an example of transmission signalswhen chirp pulses according to Variation 9 of Embodiment 1 are used;

FIG. 23 is a diagram illustrating an example of transmission signalswhen chirp pulses according to Variation 10 of Embodiment 1 are used;and

FIG. 24 is a block diagram illustrating an example configuration of aradar apparatus according to Embodiment 2.

DESCRIPTION OF EMBODIMENTS

A MIMO radar transmits signals (radar transmission waves) multiplexedby, for example, time division, frequency division, or code divisionfrom a plurality of transmit antennas (or referred to as a transmitarray antenna), receives signals (radar reflected waves) reflected froma nearby object using a plurality of receive antennas (or referred to asa receive array antenna), and separates and receives the multiplexedtransmission signals from the respective reception signals. Through theprocess described above, the MIMO radar can acquire the propagationchannel response, which is given by the product of the number oftransmit antennas and the number of receive antennas, and the receptionsignals are used as a virtual receive array to perform array signalprocessing.

In the MIMO radar, furthermore, an arrangement with appropriate elementspacing in the transmit and receive array antennas can extend theantenna aperture in a virtual fashion and improve the angularresolution.

For example, PTL 1 discloses a MIMO radar (hereinafter referred to as a“time-division multiplexing MIMO radar”) that uses, as a multiplexingtransmission method for the MIMO radar, time-division multiplexingtransmission to transmit a signal from each transmit antenna at ashifted transmission time. Time-division multiplexing transmission canbe implemented with a simpler configuration than frequency multiplexingtransmission or code multiplexing transmission. In time-divisionmultiplexing transmission, furthermore, a sufficient width of theintervals between transmission times can maintain proper orthogonalitybetween transmission signals. The time-division multiplexing MIMO radaroutputs transmission pulses, which are an example of transmissionsignals, while sequentially switching the transmit antennas in a definedperiod. The time-division multiplexing MIMO radar receives, at aplurality of receive antennas, signals that are the transmission pulsesreflected by an object, performs a correlation process between thereception signals and the transmission pulses, and then performs, forexample, spatial fast Fourier transform (FFT) processing (processing fordirection of arrival estimation of the reflected waves).

The time-division multiplexing MIMO radar sequentially switches thetransmit antennas, from which transmission signals (for example,transmission pulses or radar transmission waves) are to be transmitted,in a defined period. In time-division multiplexing transmission,therefore, it may take a longer time to complete transmission oftransmission signals from all the transmit antennas than in frequencydivision transmission or code division transmission. Accordingly, forexample, as in PTL 2, when transmission signals are transmitted from therespective transmit antennas and the Doppler frequency (i.e., therelative velocity of the target) is detected from their reception phasechanges, the time interval for observing the reception phase changes(for example, sampling interval) is increased to apply Fourier frequencyanalysis to detect the Doppler frequency. This reduces the Dopplerfrequency range over which the Doppler frequency can be detected withoutcausing aliasing (folding) (i.e., the range of detectable relativevelocities of the target).

If a reflected wave signal outside the Doppler frequency range overwhich the Doppler frequency can be detected without causing aliasing (inother words, the range of relative velocities) is assumed to come fromthe target, the radar apparatus is unable to identify whether thereflected wave signal is the aliasing component, causing the ambiguity(uncertainty) of the Doppler frequency (in other words, the relativevelocity of the target).

For example, when the radar apparatus transmits transmission signals(transmission pulses) while sequentially switching Nt transmit antennasin period Tr, it takes a transmission time given by Tr×Nt to completetransmission of the transmission signals from all the transmit antennas.As a result of repeating this time-division multiplexing transmissionoperation N_(c) times and applying Fourier frequency analysis to detectthe Doppler frequency, the Doppler frequency range over which theDoppler frequency can be detected without causing aliasing is given by±1/(2Tr×Nt) from the sampling theorem. Accordingly, the Dopplerfrequency range over which the Doppler frequency can be detected withoutcausing aliasing decreases as the number Nt of transmit antennasincreases, and the ambiguity of the Doppler frequency is likely to occureven for lower relative velocities.

In the time-division multiplexing MIMO radar, the ambiguity of theDoppler frequency, described above, is likely to occur for lowerrelative velocities. In the following, focus is on a method forsimultaneously transmitting transmission signals from a plurality oftransmit antennas in a multiplexed manner, as an example.

Examples of the method for simultaneously transmitting transmissionsignals from a plurality of transmit antennas in a multiplexed mannerinclude a method (hereinafter referred to as Doppler multiplexingtransmission) for transmitting signals so that a plurality oftransmission signals can be separated in the Doppler frequency domain onthe receiving side (see, for example, NPL 3).

In Doppler multiplexing transmission, on the transmitting side,transmission signals are simultaneously transmitted from a plurality oftransmit antennas in such a manner that, for example, with respect to atransmission signal to be transmitted from a reference transmit antenna,transmission signals to be transmitted from transmit antennas differentfrom the reference transmit antenna are given Doppler shift amountsgreater than the Doppler frequency bandwidth of reception signals. InDoppler multiplexing transmission, on the receiving side, filtering isperformed in the Doppler frequency domain to separate and receive thetransmission signals transmitted from the respective transmit antennas.

In Doppler multiplexing transmission, simultaneous transmission oftransmission signals from a plurality of transmit antennas can reducethe time interval for observing reception phase changes to apply Fourierfrequency analysis to detect the Doppler frequency (or relativevelocity), compared with time-division multiplexing transmission. InDoppler multiplexing transmission, however, since filtering is performedin the Doppler frequency domain to separate the transmission signals ofthe respective transmit antennas, the effective Doppler frequencybandwidth per transmission signal is restricted.

For example, Doppler multiplexing transmission in which a radarapparatus transmits transmission signals from Nt transmit antennas inperiod Tr will be described. As a result of repeating this Dopplermultiplexing transmission operation N_(c) times and applying Fourierfrequency analysis to detect the Doppler frequency (or relativevelocity), the Doppler frequency range over which the Doppler frequencycan be detected without causing aliasing is given by ±1/(2×Tr) from thesampling theorem. That is, in Doppler multiplexing transmission, theDoppler frequency range over which the Doppler frequency can be detectedwithout causing aliasing is increased by Nt times compared withtime-division multiplexing transmission (for example, ±1/(2Tr×Nt)).

In Doppler multiplexing transmission, as described above, filtering isperformed in the Doppler frequency domain to separate transmissionsignals. Accordingly, the effective Doppler frequency bandwidth pertransmission signal is restricted to 1/(Tr×Nt), and thus a Dopplerfrequency range similar to that in time-division multiplexingtransmission is obtained. In Doppler multiplexing transmission,furthermore, in a Doppler frequency band exceeding the effective Dopplerfrequency range per transmission signal, interference with signals inthe Doppler frequency band of any other transmission signal differentfrom the transmission signal may lead to failure to correctly separatethe transmission signals.

Accordingly, an exemplary embodiment of the present disclosure describesa method for extending the range of Doppler frequencies at whichambiguity does not occur in Doppler multiplexing transmission. With thismethod, a radar apparatus according to an exemplary embodiment of thepresent disclosure can improve target-object sensing accuracy over awider Doppler frequency range.

Embodiments of the present disclosure will be described in detail withreference to the drawings. In the embodiments, the same constituentelements are identified with the same numerals, and a descriptionthereof is omitted because of redundancy.

The following describes a configuration of a radar apparatus (in otherwords, MIMO radar configuration) having a transmitting branch in whichmultiplexed different transmission signals are simultaneously sent froma plurality of transmit antennas, and a receiving branch in which thetransmission signals are separated and subjected to receptionprocessing.

The following also describes, as an example, a configuration of a radarscheme (also referred to as, for example, chirp pulse transmission (fastchirp modulation)) that uses frequency-modulated pulse waves such aschirp pulses. Note that the modulation scheme is not limited to that forfrequency modulation. For example, an exemplary embodiment of thepresent disclosure is also applicable to a radar scheme that uses apulse compression radar configured to transmit a pulse train afterperforming phase modulation or amplitude modulation.

Further, the radar apparatus performs Doppler multiplexing transmission.In addition, in Doppler multiplexing transmission, the radar apparatusperforms coding (for example, code division multiplexing (CDM)) onsignals (hereinafter referred to as “Doppler-multiplexed transmissionsignals”) with different phase rotations (in other words, phase shifts)applied, the number of which corresponds to the number of Dopplermultiplexes, and transmits the coded signals in a multiplexed manner(hereinafter referred to as “coded Doppler multiplexing”).

[Configuration of Radar Apparatus]

FIG. 1 is a block diagram illustrating an example configuration of radarapparatus 10 according to this embodiment.

Radar apparatus 10 includes radar transmitter (transmitting branch) 100and radar receiver (receiving branch) 200.

Radar transmitter 100 generates radar signals (radar transmissionsignals) and transmits the radar transmission signals in a definedtransmission period using a transmit array antenna made up of plural(for example, Nt) transmit antennas 108.

Radar receiver 200 receives reflected wave signals, which are radartransmission signals reflected by a target object (target) (notillustrated), using a receive array antenna made up of plural receiveantennas 202-1 to 202-Na. Radar receiver 200 performs signal processingon the reflected wave signals received by respective receive antennas202 to, for example, detect the presence or absence of the target objector estimate the distance of arrival, the Doppler frequency (in otherwords, relative velocity), and the direction of arrival of the reflectedwave signals, and outputs information about the estimation results (inother words, position measurement information).

The target object is an object to be detected by radar apparatus 10.Examples of the target object include vehicles (including four-wheel andtwo-wheel vehicles), a person, and a block or a curb.

[Configuration of Radar Transmitter 100]

Radar transmitter 100 includes radar transmission signal generator 101,phase rotation amount setter 104, phase rotators 107, and transmitantennas 108.

Radar transmission signal generator 101 generates a radar transmissionsignal. Radar transmission signal generator 101 includes, for example,modulated signal emitter 102 and voltage controlled oscillator (VCO)103. The following describes each of the components of radartransmission signal generator 101.

For example, as illustrated in FIG. 2 , modulated signal emitter 102periodically emits sawtooth-shaped modulated signals. Here, the radartransmission period is represented by Tr.

VCO 103 outputs, based on the radar transmission signals (modulatedsignals) output from modulated signal emitter 102, frequency-modulatedsignals (hereinafter referred to as, for example, frequency chirpsignals or chirp signals) to phase rotators 107 and radar receiver 200(mixer section 204 described below).

Phase rotation amount setter 104 sets phase rotation amounts for phaserotators 107 (in other words, phase rotation amounts corresponding tocoded Doppler multiplexing transmission). Phase rotation amount setter104 includes, for example, Doppler shift setter 105 and coder 106.

Doppler shift setter 105 sets, for example, a phase rotation amountcorresponding to a Doppler shift amount to be applied to each radartransmission signal (for example, chirp signal).

Coder 106 sets a phase rotation amount corresponding to coding. Coder106 calculates phase rotation amounts for phase rotators 107, based on,for example, the phase rotation amounts output from Doppler shift setter105 and the phase rotation amount corresponding to coding, and outputsthe phase rotation amounts to phase rotators 107. Further, coder 106outputs, for example, information about code sequences used for coding(for example, elements of orthogonal code sequences) to radar receiver200 (for example, output switching section 209).

Phase rotators 107 applies the phase rotation amounts input from coder106 to the chirp signals input from VCO 103 and outputs signalssubjected to phase rotation to transmit antennas 108. For example, eachphase rotator 107 includes a phase shifter, a phase modulator, and so on(not illustrated). The output signals of phase rotators 107 areamplified to defined transmission power and are then radiated into aspace from the respective transmit antennas 108. In other words, phaserotation amounts corresponding to Doppler shift amounts and orthogonalcode sequences are applied to radar transmission signals, which are thentransmitted from plural transmit antennas 108 in a multiplexed manner.

Next, an example method for setting phase rotation amounts using phaserotation amount setter 104 will be described.

Doppler shift setter 105 sets phase rotation amount ϕ_(ndm) for applyingDoppler shift amount DOP_(ndm) and outputs phase rotation amount ϕ_(ndm)to coder 106. Here, ndm=1, . . . , N_(DM). N_(DM) denotes a set numberof different Doppler shift amounts and is hereinafter referred to as the“number of Doppler multiplexes”.

In Radar apparatus 10, since coding performed by coder 106 is also used,the number of Doppler multiplexes N_(DM) may be set to be smaller thanthe number Nt of transmit antennas 108 used for multiplexingtransmission. The number of Doppler multiplexes N_(DM) is greater thanor equal to 2.

Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(DM) are assigneddifferent phase rotation amounts by, for example, dividing a phaserotation range greater than or equal to 0 and less than 2π. For example,phase rotation amount ϕ_(ndm) for applying Doppler shift amountDOP_(ndm) is assigned, as given by following equation 1. In thefollowing, the angle is expressed in radian.

$\begin{matrix}\lbrack 1\rbrack &  \\{\phi_{ndm} = \frac{2{\pi( {{ndm} - 1} )}}{N_{DM}}} & ( {{Equation}1} )\end{matrix}$

In equation 1, for example, in a case where the number of Dopplermultiplexes N_(DM) is equal to 2, phase rotation amount ϕ₁ for applyingDoppler shift amount DOP₁ is equal to 0, and phase rotation amount ϕ₂for applying Doppler shift amount DOP₂ is equal to π. Likewise, inequation 1, for example, in a case where the number of Dopplermultiplexes N_(DM) is equal to 4, phase rotation amount ϕ₁ for applyingDoppler shift amount DOP₁ is equal to 0, phase rotation amount ϕ₂ forapplying Doppler shift amount DOP₂ is equal to π/2, phase rotationamount ϕ₃ for applying Doppler shift amount DOP₃ is equal to ϕ, andphase rotation amount ϕ₄ for applying Doppler shift amount DOP₄ is equalto 3π/2. In other words, intervals of phase rotation amounts ϕ_(ndm) forapplying Doppler shift amounts DOP_(ndm) are equal.

The assignment of phase rotation amounts for applying Doppler shiftamounts DOP₁, DOP₂, . . . , and DOP_(DM) is not limited to that in thisassignment method. For example, the assignment of phase rotation amountsgiven by equation 1 may be shifted. For example, phase rotation amountsmay be assigned such that ϕ_(ndm)=2π(ndm)/N_(DM). Alternatively, phaserotation amounts ϕ₁, ϕ₂, . . . , and ϕ_(DM) may be randomly assigned toDoppler shift amounts DOP₁, DOP₂, . . . , and DOP_(DM) using aphase-rotation-amount assignment table.

Coder 106 sets a phase rotation amount based on one or a plurality oforthogonal code sequences less than or equal to N_(CM) for each of phaserotation amounts ϕ₁, . . . , and ϕ_(NDM) for applying N_(DM) Dopplershift amounts output from Doppler shift setter 105. Further, coder 106sets phase rotation amounts based on both the Doppler shift amounts andthe orthogonal code sequences, that is, “coded Doppler phase rotationamounts” for generating coded Doppler multiplexed signals, and outputsthe phase rotation amounts to phase rotators 107.

The following describes an example of the operation of coder 106.

For example, coder 106 uses N_(CM) orthogonal code sequences, the numberof which is equal to the number of codes (in other words, the number ofcode multiplexes) with code length Loc.

In the following, N_(CM) orthogonal code sequences with code length Locare represented by Code_(ncm)={OC_(ncm)(1), OC_(ncm)(2), . . . ,OC_(ncm)(Loc)}. OC_(ncm)(noc) denotes the noc-th code element in thencm-th orthogonal code sequence Code_(ncm). Here, noc denotes the indexof a code element, and noc=1, . . . , Loc.

The orthogonal code sequences used in coder 106 are, for example, codesthat are orthogonal (uncorrelated) to each other. For example,orthogonal code sequences may be Walsh-Hadamard codes. In this case,code length Loc used to generate N_(CM) orthogonal code sequences, thenumber of which is equal to the number of codes, is given by followingequation 2.

[2]

Loc=2^(ceil[log) ² ^((N) ^(cm) ^()])  (Equation 2)

Here, ceil[x] denotes the operator (ceiling function) that outputs aminimum integer greater than or equal to real number x.

For example, in a case where N_(CM)=2, code length Loc of Walsh-Hadamardcodes is equal to 2, and orthogonal code sequences are represented byCode₁={1, 1} and Code₂={1, −1}. When a code element in an orthogonalcode sequence is 1, 1=exp(j0), with the phase thereof being 0. When acode element in an orthogonal code sequence is −1, −1=exp(jπ), with thephase thereof being π.

Further, for example, in a case where N_(CM)=4, code length Loc is equalto 4, and orthogonal code sequences are represented by Code₁={1, 1, 1,1}, Code₂={1, −1, 1, −1}, Code₃={1, 1, −1, −1}, and Code₄={1, −1, −1,1}.

Code elements in an orthogonal code sequence are not limited to realnumbers and may include complex number values. For example, anorthogonal code sequence Code_(ncm) given by following equation 3 may beused. Here, ncm=1, . . . , N_(CM). In this case, a code length used togenerate N_(CM) orthogonal code sequences, the number of which is equalto the number of codes, is represented by Loc=N_(CM).

$\begin{matrix}\lbrack 3\rbrack &  \\{{Code}_{ncm} = \{ {1,{\exp\lbrack {j\frac{2\pi}{N_{CM}}( {{ncm} - 1} )} \rbrack},{\exp\lbrack {j\frac{2\pi}{N_{CM}}2( {{ncm} - 1} )} \rbrack},{\ldots{\exp\lbrack {j\frac{2\pi}{N_{CM}}( {N_{CM} - 1} )( {{ncm} - 1} )} \rbrack}}} \}} & ( {{Equation}3} )\end{matrix}$

For example, in a case where N_(CM)=3, code length Loc is equal to 3(=N_(CM)), and coder 106 generates orthogonal code sequences representedby Code₁={1, 1, 1}, Code₂={1, exp(j2π/3), exp(j4π/3)}, and Code₃={1,exp(−j2π/3), exp(−j4π/3)}.

For example, in a case where N_(CM)=4, code length Loc is equal to 4(=N_(CM)), and coder 106 generates orthogonal code sequences representedby Code₁={1, 1, 1, 1}, Code₂={1, j, −1, −j}, Code₃={1,−1,1,−1}, andCode₄={1, −j, −1, j}. Here, j is the imaginary unit.

In Coder 106, the number of code multiplexes (hereinafter referred to asthe number of coded Doppler multiplexes) for coding a Dopplermultiplexed signal that uses the ndm-th Doppler shift amount DOP_(ndm)output from Doppler shift setter 105 is represented by“N_(DOP_CODE)(ndm)”. Here, ndm=1, . . . , N_(DM).

Coder 106 sets the number of coded Doppler multiplexes N_(DOP_CODE)(ndm)so that, for example, the sum of the numbers of coded Dopplermultiplexes N_(DOP_CODE)(1), N_(DOP_CODE)(2), . . . , andN_(DOP_CODE)(N_(DM)) for coding Doppler multiplexed signals is equal tothe number Nt of transmit antennas 108 used for multiplexingtransmission. In other words, coder 106 sets the number of coded Dopplermultiplexes N_(DOP_CODE)(ndm) so as to satisfy following equation 4.This enables radar apparatus 10 to perform multiplexing transmission inthe Doppler domain and the code domain (hereinafter referred to as codedDoppler multiplexing transmission) using Nt transmit antennas 108.

$\begin{matrix}\lbrack 4\rbrack &  \\{{\sum\limits_{{ndm} = 1}^{N_{DM}}{N_{DOP\_ CODE}( {ndm} )}} = {Nt}} & ( {{Equation}4} )\end{matrix}$

Here, coder 106 sets, for example, the numbers of coded Dopplermultiplexes N_(DOP_CODE)(1), N_(DOP_CODE)(2), . . . , andN_(DOP_CODE)(N_(DM)) so as to include different numbers of coded Dopplermultiplexes in the range greater than or equal to 1 and less than orequal to N_(CM). For example, coder 106 sets the numbers of codedDoppler multiplexes such that not all of the numbers of coded Dopplermultiplexes are set to N_(CM), which is equal to the number of codes,but at least one of the numbers of coded Doppler multiplexes is set tobe smaller than N_(CM). In other words, coder 106 sets the numbers ofcoded Doppler multiplexes for Doppler multiplexed signals to benon-uniform. With this setting, for example, radar apparatus 10 canindividually separate and receive signals transmitted from pluraltransmit antennas 108 in a coded Doppler multiplexed manner throughreception processing described below.

Coder 106 sets, in the m-th transmission period Tr, coded Doppler phaserotation amount ψ_(ndop_code(ndm), ndm)(m) given by following equation 5for phase rotation amount ϕ_(ndm) for applying the ndm-th Doppler shiftamount DOP_(ndm) and outputs coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm)(m) to phase rotator 107.

$\begin{matrix}\lbrack 5\rbrack &  \\{{\psi_{{{ndop\_ code}{({ndm})}},{ndm}}(m)} = {{{{floor}\lbrack \frac{( {m - 1} )}{Loc} \rbrack} \times \phi_{ndm}} + {{angle}\lbrack {O{C_{{ndop\_ code}{({ndm})}}( {O{C\_ INDEX}} )}} \rbrack}}} & ( {{Equation}5} )\end{matrix}$

Here, the subscript “ndop_code(ndm)” represents an index less than orequal to the number of coded Doppler multiplexes N_(DOP_CODE)(ndm) forphase rotation amount ϕ_(ndm) for applying Doppler shift amountDOP_(ndm). For example, ndop_code(ndm)=1, . . . , N_(DOP_CODE)(ndm).Further, angle[x] denotes the operator that outputs a radian phase ofreal number x. For example, angle[1]=0, angle[−1]=π, angle[j]=π/2, andangle[−j]=−π/2. Further, floor[x] denotes the operator that outputs amaximum integer not greater than real number x, where j is the imaginaryunit.

For example, as given by equation 5, coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm)(m) provides a constant phase rotation amount forapplying Doppler shift amount DOP_(ndm) (for example, the first term inequation 5) for the duration of Loc transmission periods, the number ofwhich is equal to the code length used for coding, and applies acorresponding phase rotation amount to each of Loc code elementsOC_(ndop_code(ndm))(1), . . . , and OC_(ndop_code(ndm))(Loc) of codeCode_(ndop_code(ndm)) used for coding (the second term in equation 5).

Further, coder 106 outputs, for each transmission period (Tr),orthogonal code element index OC_INDEX to radar receiver 200 (outputswitching section 209 described below). OC_INDEX represents anorthogonal code element index that indicates the elements of orthogonalcode sequence Code_(ndop_code(ndm)). OC_INDEX cyclically varies in therange of 1 to Loc for each transmission period (Tr), as given byfollowing equation 6.

[6]

OC_INDEX=mod(m−1,Loc)+1  (Equation 6)

Here, mod(x, y) denotes a modulo operator and is a function that outputsthe remainder of x divided by y. Further, m=1, . . . , Nc. Nc denotesthe number of transmission periods used for radar position determination(hereinafter referred to as the “number of transmissions of radartransmission signals”). The number Nc of transmissions of radartransmission signals is set to an integer multiple (Ncode times) of Loc.For example, Nc=Loc×Ncode.

Next, an example method for setting the numbers of coded Dopplermultiplexes N_(DOP_CODE)(ndm) for Doppler multiplexed signals to benon-uniform using coder 106 will be described.

For example, coder 106 sets the number of orthogonal code sequences (inother words, the number of code multiplexes or the number of codes)N_(CM) satisfying the condition below. For example, the number oforthogonal code sequences N_(CM) and the number of Doppler multiplexesN_(DM) satisfy the following relationship for the number Nt of transmitantennas 108 used for multiplexing transmission.

(Number of orthogonal code sequences N _(CM))×(number of Dopplermultiplexes N _(DM))>number Nt of transmit antennas used formultiplexing transmission

For example, among the numbers of orthogonal code sequences N_(CM) andthe numbers of Doppler multiplexes N_(DM) satisfying the above-describedcondition, the use of a combination having a smaller value of theproduct (N_(CM)×N_(DM)) is desirable in terms of both characteristicsand complexity of circuit configuration. Note that among the numbers oforthogonal code sequences N_(CM) and the numbers of Doppler multiplexesN_(DM) satisfying the above-described condition, a combination having asmaller value of the product (N_(CM)×N_(DM)) is not restrictive, and anyother combination may be applied.

For example, in a case where Nt=3, the combination of N_(DM)=2 andN_(CM)=2 is desirable.

In this case, the assignment of Doppler shift amounts DOP₁ and DOP₂ andorthogonal codes Code₁ and Code₂ is determined in accordance with, forexample, as illustrated in FIGS. 3A and 3B, the setting ofN_(DOP_CODE)(1) and N_(DOP_CODE)(2). In FIGS. 3A and 3B, white circles(“◯”) represent Doppler shift amounts and orthogonal codes used, andcrosses (“x”) represent Doppler shift amounts and orthogonal codes notused (the same applies to the following description).

For example, FIG. 3A illustrates an example where N_(DOP_CODE)(1)=2 andN_(DOP_CODE)(2)=1, and FIG. 3B illustrates an example whereN_(DOP_CODE)(1)=1 and N_(DOP_CODE)(2)=2.

In FIGS. 3A and 3B, Code₁ is used for the Doppler shift amountcorresponding to the number of coded Doppler multiplexesN_(DOP_CODE)(ndm)=1 (for example, DOP₂ in FIG. 3A and DOP₁ in FIG. 3B),which is not restrictive. For example, in a case whereN_(DOP_CODE)(1)<N_(CM) or N_(DOP_CODE)(2)<N_(CM), as illustrated inFIGS. 4A and 4B, Code₂ may be used in place of Code₁ for the Dopplershift amount corresponding to N_(DOP_CODE)(ndm)=1 (for example, DOP₂ inFIG. 4A and DOP₁ in FIG. 4B).

For example, in a case where Nt=4 or 5, the combination of N_(DM)=3 andN_(CM)=2 or the combination of N_(DM)=2 and N_(CM)=3 is desirable.

FIGS. 5A to 5C illustrate a case where Nt=4, N_(DM)=3, and N_(CM)=2, asan example. For example, the assignment of Doppler shift amounts DOP₁,DOP₂, and DOP₃ and orthogonal codes Code₁ and Code₂ is determined inaccordance with, as illustrated in FIGS. 5A to 5C, the setting ofN_(DOP_CODE)(1), N_(DOP_CODE)(2), and N_(DOP_CODE)(3).

For example, FIG. 5A illustrates an example where N_(DOP_CODE)(1)=2,N_(DOP_CODE)(2)=1, and N_(DOP_CODE)(3)=1, FIG. 5B illustrates an examplewhere N_(DOP_CODE)(1)=1, N_(DOP_CODE)(2)=2, and N_(DOP_CODE)(3)=1, andFIG. 5C illustrates an example where N_(DOP_CODE)(1)=1,N_(DOP_CODE)(2)=1, and N_(DOP_CODE)(3)=2.

In FIGS. 5A to 5C, Code₁ is used for the Doppler shift amountcorresponding to the number of coded Doppler multiplexesN_(DOP_CODE)(ndm)=1, which is not restrictive. For example, for settingsin which the numbers of coded Doppler multiplexes are each smaller thanN_(CM), Code₂ may be used in place of Code₁, as illustrated in FIG. 6A,or both Code₁ and Code₂ may be used, as illustrated in FIG. 6B or FIG.6C.

FIGS. 7A and 7B illustrate a case where Nt=4, N_(DM)=2, and N_(CM)=3, asanother example. For example, the assignment of Doppler shift amountsDOP₁ and DOP₂ and orthogonal codes Code₁, Code₂, and Code₃ is determinedin accordance with, as illustrated in FIGS. 7A and 7B, the setting ofN_(DOP_CODE)(1) and N_(DOP_CODE)(2).

For example, FIG. 7A illustrates an example where N_(DOP_CODE)(1)=3 andN_(DOP_CODE)(2)=1, and FIG. 7B illustrates an example whereN_(DOP_CODE)(1)=1 and N_(DOP_CODE)(2)=3.

In FIGS. 7A and 7B, Code₁ is used for the Doppler shift amountcorresponding to the number of coded Doppler multiplexesN_(DOP_CODE)(ndm)=1, which is not restrictive. For example, whenN_(DOP_CODE)(1)<N_(CM) or N_(DOP_CODE)(2)<N_(CM), Code₂ may be used inplace of Code₁, as illustrated in FIG. 8A, or Code₃ may be used in placeof Code₁, as illustrated in FIG. 8B.

For example, in a case where Nt=4, N_(DM)=2, and N_(CM)=3, if, asillustrated in FIGS. 9A and 9B, N_(DOP_CODE)(1)=2 and N_(DOP_CODE)(2)=2are set, the numbers of coded Doppler multiplexes N_(DOP_CODE) areuniform for Doppler shift amounts DOP₁ and DOP₂. In this setting, forexample, it is assumed that, as illustrated in FIG. 9A, the same set ofcodes (for example, Code₁ and Code₂) is assigned to Doppler shiftamounts DOP₁ and DOP₂, or that, as illustrated in FIG. 9B, differentsets of codes are assigned to Doppler shift amounts DOP₁ and DOP₂. Ineither FIG. 9A or FIG. 9B, radar apparatus 10 is capable of identifyingsignals transmitted from plural transmit antennas 108 in a coded Dopplermultiplexed manner if the Doppler frequency range is a Doppler frequencyrange within the range of 1/N_(CM) compared with the maximum Dopplervelocity at the time of single-antenna transmission.

In this embodiment, in contrast, for example, in a case where Nt=4,N_(DM)=2, and N_(CM)=3, the numbers of coded Doppler multiplexesN_(DOP_CODE) are set to be non-uniform for Doppler shift amounts DOP₁and DOP₂, such as N_(DOP_CODE)(1)=3 and N_(DOP_CODE)(2)=1, orN_(DOP_CODE)(1)=1 and N_(DOP_CODE)(2)=3, as illustrated in FIGS. 7A and7B. In this setting, the Doppler frequency range can be equivalent to,for example, the maximum Doppler velocity at the time of single-antennatransmission (the details will be described below).

For example, in a case where Nt=6 or 7, the combination of N_(DM)=4 andN_(CM)=2 or the combination of N_(DM)=2 and N_(CM)=4 is desirable.

FIGS. 10A and 10B illustrate a case where Nt=6, N_(DM)=4, and N_(CM)=2,as an example. For example, the assignment of Doppler shift amountsDOP₁, DOP₂, DOP₃, and DOP₄ and orthogonal codes Code₁ and Code₂ isdetermined in accordance with, as illustrated in FIGS. 10A and 10B, thesetting of N_(DOP_CODE)(1), N_(DOP_CODE)(2), N_(DOP_CODE)(3), andN_(DOP_CODE)(4).

For example, FIG. 10A illustrates an example whereN_(DOP_CODE)(1)=N_(DOP_CODE)(2)=2 and N_(DOP_CODE)(3)=N_(DOP_CODE)(4)=1,and FIG. 10B illustrates an example whereN_(DOP_CODE)(1)=N_(DOP_CODE)(3)=2 and N_(DOP_CODE)(2)=N_(DOP_CODE)(4)=1.

In FIGS. 10A and 10B, Code₁ is used for the Doppler shift amountcorresponding to the number of coded Doppler multiplexesN_(DOP_CODE)(ndm)=1, which is not restrictive. For example, for settingsin which the numbers of coded Doppler multiplexes are each smaller thanN_(CM), Code₂ may be used in place of Code₁, as illustrated in FIG. 11A,or both Code₁ and Code₂ may be used, as illustrated in FIG. 11B.

For example, as illustrated in FIGS. 10A and 10B, in a case where Nt=6,N_(DM)=4, and N_(CM)=2, there are two Doppler shift amounts that do notuse all the codes. Further, for example, among N_(DM)=4, forcombinations of Doppler shift amounts that do not use all the codes,there are six combinations (=₄C₂) of two Doppler shift amounts selectedfrom four Doppler shift amounts, and in each combination, there are fourcombinations (=N_(CM)×N_(CM)) of codes used. Accordingly, in a casewhere Nt=6, N_(DM)=4, and N_(CM)=2, there is a total of 24 combinationsof Doppler shift amounts DOP and orthogonal codes Code that areassigned.

Likewise, for example, in a case where Nt=8, the combination of N_(DM)=3and N_(CM)=3 or the combination of n_(DM)=5 and N_(CM)=2 is desirable.For example, in a case where Nt=9, the combination of N_(DM)=5 andN_(CM)=2 is desirable. For example, in a case where Nt=10, thecombination of N_(DM)=6 and N_(CM)=2 or the combination of N_(DM)=4 andN_(CM)=3 is desirable. The number Nt of transmit antennas 108 is notlimited to that in the example described above, and an exemplaryembodiment of the present disclosure is also applicable to Nt=11 ormore.

Next, an example of how coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm)(m) is set will be described.

For example, a description will be given of a case where in coder 106,the number of transmit antennas used for multiplexing transmission Nt=3,the number of Doppler multiplexes N_(DM)=2, N_(CM)=2, and orthogonalcode sequences Code₁={1, 1} and Code₂={1, −1} with code length Loc=2 areused. In this case, for example, if the numbers of coded Dopplermultiplexes are set such that N_(DOP_CODE)(1)=1 and N_(DOP_CODE)(2)=2,coder 106 sets coded Doppler phase rotation amounts ψ_(1, 1)(m),ψ_(1, 2)(m), and ψ_(2, 2)(m) given by following equations 7 to 9 andoutputs coded Doppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m),and ψ_(2, 2)(m) to phase rotators 107.

[7]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }={0,0,ϕ₁,ϕ₁,2ϕ₁,2ϕ₁,3ϕ₁,3ϕ₁, . . . }  (Equation 7)

[8]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }={0,0,ϕ₂,ϕ₂,2ϕ₂,2ϕ₂,3ϕ₂,3ϕ₂, . . . }  (Equation 8)

[9]

{ψ_(2,2)(1),ψ_(2,2)(2),ψ_(2,2)(3),ψ_(2,2)(4),ψ_(2,2)(5),ψ_(2,2)(6),ψ_(2,2)(7),ψ_(2,2)(8),. . . }={0,π,ϕ₂,ϕ₂+π,2ϕ₂,2ϕ₂+π,3ϕ₂,3ϕ₂+π, . . . }  (Equation 9)

Here, as an example, the phase rotation amount for applying Dopplershift amount DOP_(ndm) is given by ϕ_(ndm)=2π(ndm−1)/N_(DM) in equation1, and phase rotation amount ϕ₁ for applying Doppler shift amount DOP₁,which is equal to 0, and phase rotation amount ϕ₂ for applying Dopplershift amount DOP₂, which is equal to π, are used. In this case, coder106 sets coded Doppler phase rotation amounts Φ_(1, 1)(m), ψ_(1, 2)(m),and ψ_(2, 2)(m) given by following equations 10 to 12 and outputs codedDoppler phase rotation amounts ψ_(1, 1)(m), Φ_(1, 2)(m), and ψ_(2, 2)(m)to phase rotators 107. Here, m=1, . . . , Nc. Here, a modulo operationfor 2π is performed, and results are expressed in radians ranging from 0or more to less than 2π (the same applies to the following description).

[10]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }={0,0,0,0,0,0,0,0, . . . }  (Equation 10)

[11]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }={0,0,π,π,0,0,π,π, . . . }  (Equation 11)

[12]

{ψ_(2,2)(1),ψ_(2,2)(2),ψ_(2,2)(3),ψ_(2,2)(4),ψ_(2,2)(5),ψ_(2,2)(6),ψ_(2,2)(7),ψ_(2,2)(8),. . . }={0,π,π,0,0,π,π,0, . . . }  (Equation 12)

As given by equations 10 to 12, when a phase rotation amount is set toϕ_(ndm)=2π(ndm−1)/N_(DM), into which 2π is equally divided, codedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m)are changed in transmission periods given by N_(DM)×N_(CM)=2×2=4.

As another example, the phase rotation amount for applying Doppler shiftamount DOP_(ndm) may be set to ϕ_(ndm)=2π(ndm)/N_(DM), and phaserotation amount ϕ₁ for applying Doppler shift amount DOP₁, which isequal to π, and phase rotation amount ϕ₂ for applying Doppler shiftamount DOP₂, which is equal to 0, may be used. In this case, coder 106sets coded Doppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), andψ_(2, 2)(m) given by following equations 13 to 15 and outputs codedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m)to phase rotators 107. Here, m=1, . . . , Nc.

[13]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }={0,0,π,π,0,0,π,π, . . . }  (Equation 13)

[14]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }={0,0,0,0,0,0,0,0, . . . }  (Equation 14)

[15]

{ψ_(2,2)(1),ψ_(2,2)(2),ψ_(2,2)(3),ψ_(2,2)(4),ψ_(2,2)(5),ψ_(2,2)(6),ψ_(2,2)(7),ψ_(2,2)(8),. . . }={0,π,0,π,0,π,0,π, . . . }  (Equation 15)

As given by equations 10 to 12 or equations 13 to 15, the number ofphases (for example, two, namely, 0 and π) used for a phase rotationamount (for example, a phase rotation amount for applying a Dopplershift amount) is smaller than the number of transmit antennas 108 usedfor multiplexing transmission, namely, Nt=3. In other words, as given byequations 10 to 12 or equations 13 to 15, the number of phases (forexample, two, namely, 0 and π) used for a phase rotation amount forapplying a Doppler shift amount is equal to the number of Doppler shiftamounts used for multiplexing transmission (in other words, the numberof Doppler multiplexes) N_(DM)=2.

Further, for example, a description will be given of a case where incoder 106, the number of transmit antennas used for multiplexingtransmission Nt=6, the number of Doppler multiplexes N_(DM)=4, N_(CM)=2,and orthogonal code sequences Code₁={1, 1} and Code₂={1, −1} with codelength Loc=2 are used. In this case, for example, if the numbers ofcoded Doppler multiplexes are set such that N_(DOP_CODE)(1)=1,N_(DOP_CODE)(2)=1, N_(DOP_CODE)(3)=2, and N_(DOP_CODE)(4)=2, coder 106sets coded Doppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m),ψ_(1, 3)(m), ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m) given byfollowing equations 16 to 21 and outputs coded Doppler phase rotationamounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m), ψ_(1, 4)(m),and ψ_(2, 4)(m) to phase rotators 107. Here, m=1, . . . , Nc.

[16]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }={0,0,ϕ₁,ϕ₁,2ϕ₁,2ϕ₁,3ϕ₁,3ϕ₁, . . . }  (Equation 16)

[17]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }={0,0,ϕ₂,ϕ₂,2ϕ₂,2ϕ₂,3ϕ₂,3ϕ₂, . . . }  (Equation 17)=

[18]

{ψ_(1,3)(1),ψ_(1,3)(2),ψ_(1,3)(3),ψ_(1,3)(4),ψ_(1,3)(5),ψ_(1,3)(6),ψ_(1,3)(7),ψ_(1,3)(8),. . . }={0,0,ϕ₃,ϕ₃,2ϕ₃,2ϕ₃,3ϕ₃,3ϕ₃, . . . }  (Equation 18)

[19]

{ψ_(2,3)(1),ψ_(2,3)(2),ψ_(2,3)(3),ψ_(2,3)(4),ψ_(2,3)(5),ψ_(2,3)(6),ψ_(2,3)(7),ψ_(2,3)(8),. . . }={0,π,ϕ₃,ϕ₃+π,2ϕ₃,2ϕ₃+π,3ϕ₃,3ϕ₃+π, . . . }  (Equation 19)

[20]

{ψ_(1,4)(1),ψ_(1,4)(2),ψ_(1,4)(3),ψ_(1,4)(4),ψ_(1,4)(5),ψ_(1,4)(6),ψ_(1,4)(7),ψ_(1,4)(8),. . . }={0,0,ϕ₄,ϕ₄,2ϕ₄,2ϕ₄,3ϕ₄,3ϕ₄, . . . }  (Equation 20)

[21]

{ψ_(2,4)(1),ψ_(2,4)(2),ψ_(2,4)(3),ψ_(2,4)(4),ψ_(2,4)(5),ψ_(2,4)(6),ψ_(2,4)(7),ψ_(2,4)(8),. . . }={0,π,ϕ₄,ϕ₄+π,2ϕ₄,2ϕ₄+π,3ϕ₄,3ϕ₄+π, . . . }  (Equation 21)

Here, as an example, the phase rotation amount for applying Dopplershift amount DOP_(ndm) is given by ϕ_(ndm)=2π(ndm−1)/N_(DM), and phaserotation amount ϕ₁ for applying Doppler shift amount DOP₁, which isequal to 0, phase rotation amount ϕ₂ for applying Doppler shift amountDOP₂, which is equal to π/2, phase rotation amount ϕ₃ for applyingDoppler shift amount DOP₃, which is equal to π, and phase rotationamount ϕ₄ for applying Doppler shift amount DOP₄, which is equal to3π/2, are used. In this case, coder 106 sets coded Doppler phaserotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m),ψ_(1, 4)(m), and ψ_(2, 4)(m) given by following equations 22 to 27 andoutputs coded Doppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m),ψ_(1, 3)(m), ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m) to phase rotators107. Here, m=1, . . . , Nc.

$\begin{matrix}\lbrack 22\rbrack &  \\{\{ {{\psi_{1,1}(1)},{\psi_{1,1}(2)},{\psi_{1,1}(3)},{\psi_{1,1}(4)},{\psi_{1,1}(5)},{\psi_{1,1}(6)},{\psi_{1,1}(7)},{\psi_{1,1}(8)},\ldots} \} = \{ {0,0,0,0,0,0,0,0,\ldots} \}} & ( {{Equation}22} )\end{matrix}$ $\begin{matrix}\lbrack 23\rbrack &  \\{\{ {{\psi_{1,2}(1)},{\psi_{1,2}(2)},{\psi_{1,2}(3)},{\psi_{1,2}(4)},{\psi_{1,2}(5)},{\psi_{1,2}(6)},{\psi_{1,2}(7)},{\psi_{1,2}(8)},\ldots} \} = \{ {0,0,\frac{\pi}{2},\frac{\pi}{2},\pi,\pi,\frac{3\pi}{2},\frac{3\pi}{2},\ldots} \}} & ( {{Equation}23} )\end{matrix}$ $\begin{matrix}\lbrack 24\rbrack &  \\{\{ {{\psi_{1,3}(1)},{\psi_{1,3}(2)},{\psi_{1,3}(3)},{\psi_{1,3}(4)},{\psi_{1,3}(5)},{\psi_{1,3}(6)},{\psi_{1,3}(7)},{\psi_{1,3}(8)},\ldots} \} = \{ {0,0,\pi,\pi,0,0,\pi,\pi,\ldots} \}} & ( {{Equation}24} )\end{matrix}$ $\begin{matrix}\lbrack 25\rbrack &  \\{\{ {{\psi_{2,3}(1)},{\psi_{2,3}(2)},{\psi_{2,3}(3)},{\psi_{2,3}(4)},{\psi_{2,3}(5)},{\psi_{2,3}(6)},{\psi_{2,3}(7)},{\psi_{2,3}(8)},\ldots} \} = \{ {0,\pi,\pi,0,0,\pi,\pi,0,\ldots} \}} & ( {{Equation}25} )\end{matrix}$ $\begin{matrix}\lbrack 26\rbrack &  \\{\{ {{\psi_{1,4}(1)},{\psi_{1,4}(2)},{\psi_{1,4}(3)},{\psi_{1,4}(4)},{\psi_{1,4}(5)},{\psi_{1,4}(6)},{\psi_{1,4}(7)},{\psi_{1,4}(8)},\ldots} \} = \{ {0,0,\frac{3\pi}{2},\frac{3\pi}{2},\pi,\pi,\frac{\pi}{2},\frac{\pi}{2},\ldots} \}} & ( {{Equation}26} )\end{matrix}$ $\begin{matrix}\lbrack 27\rbrack &  \\{\{ {{\psi_{2,4}(1)},{\psi_{2,4}(2)},{\psi_{2,4}(3)},{\psi_{2,4}(4)},{\psi_{2,4}(5)},{\psi_{2,4}(6)},{\psi_{2,4}(7)},{\psi_{2,4}(8)},\ldots} \} = \{ {0,\pi,\frac{3\pi}{2},\frac{\pi}{2},\pi,0,\frac{\pi}{2},\frac{3\pi}{2},\ldots} \}} & ( {{Equation}27} )\end{matrix}$

As given by equations 22 to 27, when a phase rotation amount is set toϕ_(ndm)=2π(ndm−1)/N_(DM), into which 2π is equally divided, codedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ_(1, 3)(m),ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m) are changed in transmissionperiods given by N_(DM)×N_(CM)=4×2=8.

As given by equations 22 to 27, furthermore, the number of phases (forexample, four, namely, 0, π/2, π, and 3π/2) used for a phase rotationamount (for example, a phase rotation amount for applying a Dopplershift amount) is smaller than the number of transmit antennas 108 usedfor multiplexing transmission, namely, Nt=6. In other words, as given byequations 22 to 27, the number of phases (for example, four, namely, 0,π/2, π, and 3π/2) used for a phase rotation amount for applying aDoppler shift amount is equal to the number of Doppler shift amountsused for multiplexing transmission (in other words, the number ofDoppler multiplexes) N_(DM)=4.

While the description has been given of, as an example, the setting ofphase rotation amounts in a case where the number Nt of transmitantennas 108 is equal to 3 and the number of Doppler multiplexes N_(DM)is equal to 2 and in a case where the number Nt of transmit antennas 108is equal to 6 and the number of Doppler multiplexes N_(DM) is equal to4, the number Nt of transmit antennas 108 and the number of Dopplermultiplexes N_(DM) are not limited to the values described above. Forexample, the number of phases used for a phase rotation amount may beset to be smaller than the number Nt of transmit antennas 108 used formultiplexing transmission, regardless of the number Nt of transmitantennas 108. Further, the number of phases used for a phase rotationamount for applying a Doppler shift amount may be equal to the numberN_(DM) of Doppler shift amounts used for multiplexing transmission.

The foregoing description has been given of a method for setting phaserotation amounts using phase rotation amount setter 104.

In FIG. 1 , each phase rotator 107 applies a phase rotation amount to achirp signal output from radar transmission signal generator 101, foreach transmission period Tr, based on the coded Doppler phase rotationamount ψ_(ndop_code(ndm), ndm)(m) set by phase rotation amount setter104. Here, ndm=1, . . . , N_(DM), and ndop_code(ndm)=1, . . . ,N_(DOP_CODE)(ndm).

The sum of the numbers of coded Doppler multiplexes N_(DOP_CODE)(1),N_(DOP_CODE)(2), . . . , and N_(DOP_CODE)(N_(DM)) is set to be equal tothe number Nt of transmit antennas 108, and Nt coded Doppler phaserotation amounts are respectively input to Nt phase rotators 107.

Each of Nt phase rotators 107 applies coded Doppler phase rotationamount ψ_(ndop_code(ndm), ndm)(m) input thereto to a chirp signal outputfrom radar transmission signal generator 101 for each transmissionperiod Tr. The outputs of Nt phase rotators 107 (referred to as, forexample, coded Doppler multiplexed signals) are amplified to definedtransmission power and are then radiated into a space from Nt transmitantennas 108 in a transmit array antenna section.

In the following, phase rotator 107 that applies coded Doppler phaserotation amount ψ_(ndop_code(ndm), ndm)(m) is as represented by “phaserotator PROT #[ndop_code(ndm), ndm]”. Likewise, transmit antenna 108that radiates the output of phase rotator PROT #[ndop_code(ndm), ndm]into a space is represented by “transmit antenna Tx #[ndop_code(ndm),ndm]”. Here, ndm=1, . . . , N_(DM), and ndop_code(ndm)=1, . . . ,N_(DOP_CODE)(ndm).

For example, a description will be given of a case where when the numberof transmit antennas used for multiplexing transmission Nt=3, the numberof Doppler multiplexes N_(DM)=2, N_(CM)4=2, orthogonal code sequencesCode₁={1, 1} and Code₂={1, −1} with code length Loc=2 are used, and thenumbers of coded Doppler multiplexes are set such that N_(DOP_CODE)(1)=1and N_(DOP_CODE)(2)=2. In this case, coded Doppler phase rotationamounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) are output from coder106 to phase rotators 107 in respective transmission periods.

For example, phase rotator PROT #[1, 1] applies, for each transmissionperiod, phase rotation amount ψ_(1, 1)(m) as given by following equation28 to a chirp signal generated by radar transmission signal generator101 for each transmission period. The output of phase rotator PROT #[1,1] is output from transmit antenna Tx #[1, 1]. Here, cp(t) denotes achirp signal for each transmission period.

[28]

exp[jψ _(1,1)(1)]cp(t),exp[jψ _(1,1)(2)]cp(t),exp[jψ _(1,1)(3)]cp(t), .. . ,exp[jψ _(1,1)(Nc)]cp(t)   (Equation 28)

Likewise, phase rotator PROT #[1, 2] applies, for each transmissionperiod, phase rotation amount ψ_(1, 2)(m) as given by following equation29 to a chirp signal generated by radar transmission signal generator101 for each transmission period. The output of phase rotator PROT #[1,2] is output from transmit antenna Tx #[1, 2].

[29]

exp[jψ _(1,2)(1)]cp(t),exp[jψ _(1,2)(2)]cp(t),exp[jψ _(1,2)(3)]cp(t), .. . ,exp[jψ _(1,2)(Nc)]cp(t)  (Equation 29)

Likewise, phase rotator PROT #[2, 2] applies, for each transmissionperiod, phase rotation amount ψ_(2, 2)(m) as given by following equation30 to a chirp signal generated by radar transmission signal generator101 for each transmission period. The output of phase rotator PROT #[2,2] is output from transmit antenna Tx #[2, 2].

[30]

exp[jψ _(2,2)(1)]cp(t),exp[jψ _(2,2)(2)]cp(t),exp[jψ _(2,2)(3)]cp(t), .. . ,exp[jψ _(2,2)(Nc)]cp(t)   (Equation 30)

The foregoing description has been given of an example of how codedDoppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m) is set.

In this embodiment, accordingly, plural transmit antennas 108 areassociated with combinations (in other words, assignment) of Dopplershift amounts DOP_(ndm) and orthogonal code sequences Code_(ncm) suchthat in each of the combinations, at least one of Doppler shift amountDOP_(ndm) or orthogonal code sequence Code_(ncm) is different. In thisembodiment, furthermore, the number of multiplexes of orthogonal codesequence Code_(ncm) (in other words, the number of coded Dopplermultiplexes N_(DOP_CODE)(ndm)) corresponding to each Doppler shiftamount DOP_(ndm) in combinations of Doppler shift amounts DOP_(ndm) andorthogonal code sequences Code_(ncm) is different.

For example, in this embodiment, as illustrated in FIGS. 3A and 3B, Nttransmit antennas 108 include at least plural transmit antennas 108 fromwhich transmission signals that are code-multiplexed using differentorthogonal code sequences are transmitted, and at least one transmitantenna 108 from which a transmission signal that is notcode-multiplexed is transmitted. In other words, radar transmissionsignals transmitted from radar transmitter 100 include at least a codedDoppler multiplexed signal for which the number of coded Dopplermultiplexes N_(DOP_CODE)(ndm) is set to the number of codes N_(CM), anda coded Doppler multiplexed signal for which the number of coded Dopplermultiplexes N_(DOP_CODE)(ndm) is set to be smaller than the number ofcodes N_(CM).

[Configuration of Radar Receiver 200]

In FIG. 1 , radar receiver 200 includes Na receive antennas 202, whichconstitute an array antenna. Radar receiver 200 further includes Naantenna channel processors 201-1 to 201-Na, constant false alarm rate(CFAR) section 211, coded Doppler demultiplexer 212, and directionestimator 213.

Each receive antenna 202 receives a reflected wave signal that is aradar transmission signal reflected from a target object (target), andoutputs the received reflected wave signal to the corresponding one ofantenna channel processors 201 as a reception signal.

Each antenna channel processor 201 includes receiving radio section 203and signal processor 206.

Receiving radio section 203 includes mixer section 204 and low passfilter (LPF) 205. Receiving radio section 203 mixes, using mixer section204, a chirp signal input from radar transmission signal generator 101,which is a transmission signal, with the received reflected wave signal,and passes the resulting mixed signal through LPF 205. As a result, abeat signal having a frequency corresponding to the delay time of thereflected wave signal is acquired. For example, as illustrated in FIG. 2, the difference frequency between the frequency of a transmission chirpsignal (transmission-frequency-modulated wave) and the frequency of areception chirp signal (reception-frequency-modulated wave) is obtainedas the beat frequency.

In each antenna channel processor 201-z (where z is equal to any of 1 toNa), signal processor 206 includes analog-to-digital (AD) converter 207,beat frequency analyzer 208, output switching section 209, and Doppleranalyzers 210.

The signal (for example, beat signal) output from LPF 205 is passed tosignal processor 206 and is converted into a discretely sampled data byAD converter 207.

Beat frequency analyzer 208 performs FFT processing of N_(data) piecesof discretely sampled data, which are obtained in a defined time range(range gate), for each transmission period Tr. As a result, signalprocessor 206 outputs a frequency spectrum in which a peak appears at abeat frequency corresponding to the delay time of the reflected wavesignal (radar reflected wave). In the FFT processing, for example, beatfrequency analyzer 208 may perform multiplication by a window functioncoefficient such as of the Han window or the Hamming window. The use ofa window function coefficient can suppress sidelobes around the beatfrequency peak.

Here, a beat frequency response obtained by the m-th chirp pulsetransmission, which is output from beat frequency analyzer 208 in thez-th signal processor 206, is represented by RET_(z)(f_(b), m). Here, fbdenotes the beat frequency index and corresponds to FFT index (binnumber). For example, f_(b)=0, . . . , N_(data)/2, z=0, Na, and m=1, . .. , N_(C). A beat frequency having smaller beat frequency index f_(b)indicates that the delay time of a reflected wave signal is shorter (inother words, the distance to the target object is shorter).

Beat frequency index f_(b) can be converted into distance informationR(f_(b)) using following equation 31. For this reason, beat frequencyindex f_(b) is hereinafter referred to as the “distance index f_(b)”.

$\begin{matrix}\lbrack 31\rbrack &  \\{{R( f_{b} )} = {\frac{C_{0}}{2B_{w}}f_{b}}} & ( {{Equation}31} )\end{matrix}$

Here, B_(w) denotes a frequency-modulation bandwidth in a range gate fora chirp signal, and C₀ denotes the speed of light.

Output switching section 209 selectively switches and outputs the outputof beat frequency analyzer 208 for each transmission period to theOC_INDEX-th Doppler analyzer 210 among Loc Doppler analyzers 210, basedon orthogonal code element index OC_INDEX output from coder 106 in phaserotation amount setter 104. In other words, output switching section 209selects the OC_INDEX-th Doppler analyzer 210 in the m-th transmissionperiod Tr.

Signal processor 206 includes Loc Doppler analyzers 210-1 to 210-Loc.For example, data is input to the noc-th Doppler analyzer 210 for everyLoc transmission periods (Loc×Tr) by using output switching section 209.Accordingly, the noc-th Doppler analyzer 210 performs Doppler analysisfor each distance index fb using data of Ncode transmission periodsamong Nc transmission periods (for example, beat frequency responseRFT_(z)(f_(b), m) output from beat frequency analyzer 208). Here, nocdenotes the index of a code element, and noc=1, . . . , Loc.

For example, when Ncode is a power of 2, FFT processing is applicable inDoppler analysis. In this case, the FFT size is Ncode, and a maximumDoppler frequency without causing aliasing derived from the samplingtheorem is ±1/(2Loc×Tr). Further, the Doppler frequency interval forDoppler frequency index f_(s) is 1/(Ncode×Loc×Tr), and the range ofDoppler frequency index f_(s) is given by f_(s)=−Ncode/2, . . . , 0, . .. , Ncode/2−1.

The following description will be given of a case where Ncode is a powerof 2, as an example. When Ncode is not a power of 2, for example, datawith zero padding can be used to perform FFT processing with a number ofdata sizes (FFT sizes) equal to a power of 2. In the FFT processing,Doppler analyzer 210 may perform multiplication by a window functioncoefficient such as of the Han window or the Hamming window. Theapplication of a window function can suppress sidelobes around the beatfrequency peak.

For example, output VFT_(z) ^(noc)(f_(b), f_(s)) of Doppler analyzers210 in the z-th signal processor 206 is given by following equation 32,where j is the imaginary unit and z=1 to Na.

$\begin{matrix}\lbrack 32\rbrack &  \\{{{VFT}_{\mathcal{z}}^{noc}( {f_{b},f_{s}} )} = {\sum\limits_{s = 0}^{N_{code} - 1}{RF{T_{\mathcal{z}}( {f_{b},{{L_{OC} \times s} + {noc}}} )}{\exp\lbrack {{- j}\frac{2\pi{sf}_{s}}{N_{code}}} \rbrack}}}} & ( {{Equation}32} )\end{matrix}$

The foregoing description has been given of the processing performed bythe components of signal processor 206.

In FIG. 1 , CFAR section 211 performs CFAR processing (in other words,adaptive threshold determination) using the outputs of Loc Doppleranalyzers 210 in each of the first to Na-th signal processors 206 andextracts distance index f_(b_cfar) and Doppler frequency indexf_(s_cfar) that provide a peak signal.

For example, CFAR section 211 performs two-dimensional CFAR processingwith the distance axis and the Doppler frequency axis (corresponding tothe relative velocity) or CFAR processing that is a combination ofone-dimensional CFAR processing operations by power addition of outputsVFT_(z) ^(noc)(f_(b), f_(s)) of Doppler analyzers 210 in the first toNa-th signal processors 206, for example, as given by following equation33. As the two-dimensional CFAR processing or the CFAR processing thatis a combination of one-dimensional CFAR processing operations, forexample, processing disclosed in NPL 2 may be applied.

$\begin{matrix}\lbrack 33\rbrack &  \\{{{PowerFT}( {f_{b},f_{s}} )} = {\sum\limits_{{\mathcal{z}} = 1}^{N_{a}}{\sum\limits_{{noc} = 1}^{L_{oc}}{❘{{VFT}_{\mathcal{z}}^{noc}( {f_{b},f_{s}} )}❘}^{2}}}} & ( {{Equation}33} )\end{matrix}$

CFAR section 211 adaptively sets a threshold and outputs distance indexf_(b_cfar) that provides received power greater than the threshold,Doppler frequency index f_(s_cfar), and received-power informationPowerFT(f_(b_cfar), f_(s_cfar)) to coded Doppler demultiplexer 212.

For example, when phase rotation amount ϕ_(ndm) for applying Dopplershift amount DOP_(ndm) is determined using equation 1, intervals ΔFD ofDoppler shift amounts in the Doppler frequency domain, which are outputfrom Doppler analyzers 210, are equal, where ΔFD=Ncode/N_(DM).Accordingly, in the outputs of Doppler analyzers 210, peaks are detectedat intervals of ΔFD for Doppler-shift multiplexed signals in the Dopplerfrequency domain. When phase rotation amount ϕ_(ndm) is determined usingequation 1, ΔFD may not sometimes be an integer depending on Ncode andN_(DM). In this case, equation 50 described below may be used to achieveΔFD having an integer value. The following describes a receptionprocessing operation using ΔFD having an integer value.

FIG. 12A illustrates an example of the outputs of Doppler analyzers 210at the distances over which reflected waves from three targets exist ina case where N_(DM)=2. For example, as illustrated in FIG. 12A, whenreflected waves from the three targets are observed at Doppler frequencyindices f1, f2, and f3, the reflected waves are also observed atrespective Doppler frequency indices (for example, f1+ΔFD, f2+ΔFD, andf3+ΔFD) at intervals of ΔFD relative to f1, f2 and f3.

Accordingly, CFAR section 211 may divide the respective outputs ofDoppler analyzers 210 into ranges at Doppler shift amount intervals ΔFDand CFAR processing (referred to as, for example, “Doppler domaincompression CFAR processing”) after, as given by following equation 34,providing power addition (referred to as, for example, “Doppler domaincompression”) for the respective ranges while matching peak positions ofDoppler-shift multiplexed signals. Here, f_(s_comp)=−Ncode/2, . . . ,−Ncode/2+ΔFD−1.

$\begin{matrix}\lbrack 34\rbrack &  \\{{{PowerFT\_ comp}( {f_{b},f_{s\_ comp}} )} = {\sum\limits_{{nfd} = 1}^{N_{DM}}{{PowerFT}( {f_{b},{f_{s\_ comp} + {( {{nfd} - 1} ) \times \Delta FD}}} )}}} & ( {{Equation}34} )\end{matrix}$

This can compress the Doppler frequency range for the CFAR processing to1/N_(DM) to reduce the amount of CFAR processing and can simplify thecircuit configuration. In addition, CFAR section 211 enables poweraddition for N_(DM) Doppler-shift multiplexed signals, resulting insignal to noise ratio (SNR) being improved by about (N_(DM))^(1/2). As aresult, the radar sensing performance of radar apparatus 10 can beimproved.

FIG. 12B illustrates an example of the outputs of Doppler analyzers 210illustrated in FIG. 12A after application of the Doppler domaincompression process given by equation 34. As illustrated in FIG. 12B, ina case where N_(DM)=2, through the Doppler domain compression process,the power component for Doppler frequency index f1 and the powercomponent for f1+ΔFD are added together and the sum is output. Likewise,as illustrated in FIG. 12B, the power component for Doppler frequencyindex f2 and the power component for f2+ΔFD are added together and thesum is output, and the power component for Doppler frequency index f2and the power component for f3+ΔFD are added together and the sum isoutput.

As a result of Doppler domain compression, the range of Dopplerfrequency index fs_comp in the Doppler frequency domain is reduced togreater than or equal to −Ncode/2, . . . , and less than or equal to−Ncode/2+ΔFD−1, and the range for the CFAR processing is compressed,leading to a reduction in the operation amount of CFAR processing. InFIGS. 12A and 12B, for example, because of power addition for thereflected waves from the three targets, the SNR of the signal componentsis improved. Note that combined power of noise components results in anSNR improvement of, for example, about (N_(DM))^(1/2).

For example, CFAR section 211, which uses Doppler domain compressionCFAR processing, adaptively sets a threshold and outputs distance indexf_(b_cfar) that provides received power greater than the threshold,Doppler frequency index f_(s_comp_cfar), and received-power informationPowerFT(f_(b_cfar), f_(s_comp_cfar)+(nfd−1)×ΔFD) for Doppler frequencyindices (f_(s_comp_cfar)+(nfd−1)×ΔFD) of N_(DM) Doppler multiplexedsignals, where nfd=1, . . . , N_(DM), to coded Doppler demultiplexer212.

Note that phase rotation amount ϕ_(ndm) for applying Doppler shiftamount DOP_(ndm) is not limited to that given in equation 1. CFARsection 211 can apply Doppler domain compression CFAR processing, forexample, if Doppler-shift multiplexed signals have phase rotationamounts ϕ_(ndm) such that peaks are detected at constant intervals inthe Doppler frequency domain output from Doppler analyzers 210.

Next, an example of the operation of coded Doppler demultiplexer 212illustrated in FIG. 1 will be described. The following describes anexample of processing performed by coded Doppler demultiplexer 212 whenCFAR section 211 uses Doppler domain compression CFAR processing.

Coded Doppler demultiplexer 212 separates signals transmitted in a codedDoppler multiplexed manner, using the output of Doppler analyzers 210,based on the outputs of CFAR section 211, namely, distance indexf_(b_cfar), Doppler frequency index f_(s_comp_cfar), and received powerinformation PowerFT(f_(b_cfar), f_(s_comp_cfar)+(nfd−1)×ΔFD) for Dopplerfrequency indices (f_(s_comp_cfar)+(nfd−1)×ΔFD) of N_(DM) Dopplermultiplexed signals, where nfd=1, . . . , N_(DM), and performsdiscrimination (in other words, also referred to as determination oridentification) of transmit antennas 108 and discrimination of Dopplerfrequencies (in other words, Doppler velocities or relative velocities).

As described above, for example, coder 106 in phase rotation amountsetter 104 does not set all of N_(DM) numbers of coded Dopplermultiplexes N_(DOP_CODE)(1), N_(DOP_CODE)(2), . . . , andN_(DOP_CODE)(N_(DM)) to N_(CM), but sets at least one of the numbers ofcoded Doppler multiplexes to a value smaller than N_(CM). For example,coded Doppler demultiplexer 212 performs (1) code separation process todetect a coded Doppler multiplexed signal for which the number of codedDoppler multiplexes is set to be smaller than N_(CM) and performs (2)discrimination of transmit antennas 108 and discrimination of Dopplerfrequencies of the target based on the detected coded Dopplermultiplexed signal for which the number of coded Doppler multiplexes isset to be smaller than N_(CM).

The following describes the processes (1) and (2) described above, whichare performed by coded Doppler demultiplexer 212.

<(1) Code Separation Process (Process for Detecting Coded DopplerMultiplexed Signal for which Number of Coded Doppler Multiplexes is Setto be Smaller than N_(CM))>

There are N_(DM) candidate correspondences between N_(DM) coded Dopplermultiplexed signals and the outputs of the respective Doppler analyzers210 for Doppler frequency indices (f_(s_comp_cfar)),(f_(s_comp_cfar)+ΔFD), (f_(s_comp_cfar)+2ΔFD), . . . , and(f_(s_comp_cfar)+(N_(DM)−1)ΔFD) of N_(DM) coded Doppler multiplexedsignals for distance index f_(b_cfar) output from CFAR section 211.

For example, if Doppler shift amount DOP_(ndm) set in Doppler shiftsetter 105 is represented by DOP₁<DOP₂< . . . <DOP_(DM-1)<DOP_(DM),there are N_(DM) candidate correspondences with cyclically shiftedelements, as follows, with consideration given to Doppler aliasing.Here, patterns of the candidate correspondences are numbered DopCase=1to N_(DM).

DopCase=1: {DOP₁, DOP₂, . . . , DOP_(DM-1), DOP_(DM)}

DopCase=2: {DOP_(DM), DOP₁, DOP₂, . . . , DOP_(DM-1)}

. . . ,

DopCase=N_(DM): {DOP₂, . . . , DOP_(DM-1), DOP_(DM), DOP₁}

For example, DopCase=1 indicates a correspondence among Doppler shiftamounts in the initial state (when the relative velocity to the targetis zero). For example, more aliasing components are included as therelative velocity of the target increases in a direction in which thedistance to the target decreases, and the resulting correspondences areassociated with DopCase=2, 3, . . . , N_(DM). In other words,DopCase=N_(DM), N_(DM-1), . . . , 2 is associated as the relativevelocity of the target increases in a direction in which the distance tothe target increases.

Here, a table indicating the position of DOP_(ndm) counting from thebeginning of each DopCase (the position (or order) of DOP_(ndm) inDopCase) can be created in advance, based on the Doppler shift amountsset in Doppler shift setter 105. In the following, DOPposi(DOP_(ndm),DopCase) denotes the operator that outputs the position of DOP_(ndm)counting from the beginning of each DopCase. For example, in theabove-described example of DopCase, DOPposi(DOP₁, 1)=1, DOPposi(DOP₁,2)=2, DOPposi(DOP₁, N_(DM))=N_(DM), DOPposi(DOP₂, 1)=2, DOPposi(DOP₂,2)=3, and DOPposi(DOP₂, N_(DM))=1.

FIG. 13 illustrates an example of the outputs of Doppler analyzers 210in a case where, as an example, N_(DM)=4, N_(CM)=2, Loc=2, the phaserotation amount for applying Doppler shift amount DOP_(ndm) is given byϕ_(ndm)=2π(ndm+1)/N_(DM), and Doppler shift amounts satisfyDOP₁<DOP₂<DOP₃<DOP₄ using phase rotation amount ϕ₁ for applying Dopplershift amount DOP₁, which is equal to π, phase rotation amount ϕ₂ forapplying Doppler shift amount DOP₂, which is equal to 3π/2, phaserotation amount ϕ₃ for applying Doppler shift amount DOP₃, which isequal to 0, and phase rotation amount ϕ₄ for applying Doppler shiftamount DOP₄, which is equal to π/2. In FIG. 13 , the horizontal axisrepresents the target Doppler frequency (f_(TARGET)), and the verticalaxis represents the output of Doppler analyzers 210.

In Doppler analyzers 210, the range of Doppler frequencies withoutcausing aliasing is greater than or equal to −1/(2×Loc×Tr) and less than1/(2×Loc×Tr), and Doppler aliasing occurs outside this range. Forexample, in a case where Loc=2, in the outputs of Doppler analyzers 210,the range of Doppler frequencies without causing aliasing is greaterthan or equal to 1/(4×Tr) and less than 1/(4×Tr).

In FIG. 13 , accordingly, the order of coded Doppler multiplexed signalscyclically changes as follows depending on Doppler frequency f_(TARGET)of the target.

DopCase=1: when −1/(2×Tr)≤f_(TARGET)<−3/(8×Tr), DOP₁<DOP₂<DOP₃<DOP₄

DopCase=2: when −3/(8×Tr)≤f_(TARGET)<−1/(4×Tr), DOP₄<DOP₁<DOP₂<DOP₃

DopCase=3: when −1/(4×Tr)≤f_(TARGET)<−1/(8×Tr), DOP₃<DOP₄<DOP₁<DOP₂

DopCase=4: when −1/(8×Tr)≤f_(TARGET)<0, DOP₂<DOP₃<DOP₄<DOP₁

DopCase=1: when 0≤f_(TARGET)<1/(8×Tr), DOP₁<DOP₂<DOP₃<DOP₄

DopCase=2: when 1/(8×Tr)≤f_(TARGET)<1/(4×Tr), DOP₄<DOP₁<DOP₂<DOP₃

DopCase=3: when 1/(4×Tr)≤f_(TARGET)<3/(8×Tr), DOP₃<DOP₄<DOP₁<DOP₂

DopCase=4: when 3/(8×Tr)≤f_(TARGET)<1/(2×Tr), DOP₂<DOP₃<DOP₄<DOP₁

Here, for Doppler frequency f_(TARGET) of the target in the range of−1/(2×Tr)≤f_(TARGET)<1/(2×Tr), the candidate patterns of the order ofcoded Doppler multiplexed signals are numbered DopCase=1 to 4 (=N_(DM)).There are 4 (=N_(DM)) candidate patterns to be associated with the orderof coded Doppler multiplexed signals.

In the outputs of Doppler analyzers 210, furthermore, the range ofDoppler frequencies without causing aliasing is greater than or equal to−1/(2×Loc×Tr) and less than 1/(2×Loc×Tr). Thus, for Doppler frequencyf_(TARGET) of the target in the range of −1/(2×Tr)≤f_(TARGET)<1/(2×Tr),the outputs of Doppler analyzers 210 include (Loc−1) occurrences ofaliasing. Therefore, for example, in the example (Loc=2) illustrated inFIG. 13 , the outputs of Doppler analyzers 210 corresponding toDopCase=1 to 4 (=N_(DM)) are output 2 (=Loc) times including the casewhere aliasing is present and the case where aliasing is absent.Accordingly, to detect Doppler frequency f_(TARGET) of the target in therange of −1/(2×Tr)≤f_(TARGET)<1/(2×Tr), for example, coded Dopplerdemultiplexer 212 performs a process of determining a DopCase andfurther determining the presence or absence of aliasing (an example ofthe process will be described below).

Coded Doppler demultiplexer 212 performs a code separation process on,for example, the outputs of Doppler analyzers 210 in the z-th signalprocessor 206 indicated by Doppler frequency indices(f_(s_comp_cfar)±(nfd−1)×ΔFD) of N_(DM) coded Doppler multiplexedsignals for distance index f_(b_cfar) output from CFAR section 211.

The code separation process may be performed on, for example, all ofnfd=1, . . . , N_(DM) for all the candidates of DopCase=1, . . . ,N_(DM). Note that coded Doppler demultiplexer 212 detects a codedDoppler multiplexed signal for which the number of coded Dopplermultiplexes is set to be smaller than N_(CM), and performsdiscrimination of transmit antennas 108 and determination of a targetDoppler frequency. Accordingly, for example, coded Doppler demultiplexer212 performs a code separation process, as given by following equation35, to reduce the operation amount of the separation process.

$\begin{matrix}\lbrack 35\rbrack &  \\{{{DeMUL}_{\mathcal{z}}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }\text{ }{\{ {{{DOPposi}( {{DOP}_{ndm1},{DopCase}} )} - 1} \} \times \Delta FD}}} )} = {\sum\limits_{{noc} = 1}^{Loc}\lbrack {O{C_{ncm}^{*}( {noc} )}{{VFT}_{\mathcal{z}}^{noc}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{ndm1},{DopCase}} )} - 1} \} \times \Delta FD}}} )} \times \text{ }\exp\{ {{- j}\frac{\begin{matrix}{2\pi( {f_{{s\_ comp}{\_ cfar}} + ( {{DopCase} - 1} )} } \\ {{\times \Delta FD} - {DOP}_{1}} )\end{matrix}}{N_{code}} \times \frac{{noc} - 1}{Loc}} \}} \rbrack}} & ( {{Equation}35} )\end{matrix}$

The superscript asterisk (*) indicates the complex conjugate operator.Further, nfd=1, . . . , N_(DM), ncm=1, . . . , N_(CM), and DopCase=1, .. . , N_(DM).

For example, when a coded Doppler multiplexed signal for which thenumber of coded Doppler multiplexes is set to be smaller than N_(CM) isDoppler shift amount “DOP_(ndm1)”, coded Doppler demultiplexer 212performs a code separation process using candidate DOPposi(DOP_(ndm1),DopCase) including Doppler shift amount DOP_(ndm1) in each DopCase.

Here, code separation signal DeMUL_(z) ^(ncm)(f_(b_cfar),f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) obtained inequation 35 represents a code separation signal that uses the ncm-thorthogonal code sequence Code_(ncm) for the output of Doppler analyzer210 in the z-th signal processor 206 for distance index f_(b_cfar) andDoppler frequency index (f_(s_comp_cfar)±(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD).

In the exp term of equation 35, since the sampling times for the outputsof Doppler analyzers 210 for each code element OC_(ncm)(noc) areshifted, phase correction is performed in accordance with Dopplerfrequency index (f_(s_comp_cfar)±(DopCase−1))×ΔFD-DOP₁).

Further, Doppler shift amount DOP_(ndm) set in Doppler shift setter 105is represented by DOP₁<DOP₂< . . . <DOP_(DM-1)<DOP_(DM), and DOP₁ fallswithin the range of f_(s_comp)=−Ncode/2, −Ncode/2+ΔFD−1 in the initialstate (when the relative velocity to the target is zero). Accordingly,for example, coded Doppler demultiplexer 212 calculates an amount ofphase correction using DOP₁ as a reference.

When a coded Doppler multiplexed signal for which the number of codedDoppler multiplexes is set to be smaller than N_(CM) is Doppler shiftamount DOP_(ndm1), signals separated based on N_(CM) orthogonal codesare obtained by equation 35 for candidate DOPposi(DOP_(ndm1), DopCase)including Doppler shift amount DOP_(ndm1), where DopCase=1, . . . ,N_(DM). Accordingly, coded Doppler demultiplexer 212 obtains the outputsof a total of N_(DM)×N_(CM) code separation signals. For example, codedDoppler demultiplexer 212 calculates code separation signals for allreceive antennas z=1, Na in accordance with equation 35 and calculatescode separation signal power sum Pow_DeMUL^(ncm)(f_(b_car),f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) using followingequation 36.

$\begin{matrix}\lbrack 36\rbrack &  \\{{{Pow\_ DeMUL}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{ndm1},{DopCase}} )} - 1} \} \times \Delta FD}}} )} = {\sum\limits_{{\mathcal{z}} = 1}^{Na}{❘{{DeMUL}_{\mathcal{z}}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{ndm1},{DopCase}} )} - 1} \} \times \Delta FD}}} )}❘}^{2}}} & ( {{Equation}36} )\end{matrix}$

Here, a coded Doppler multiplexed signal for which the number of codedDoppler multiplexes is set to be smaller than N_(CM) does not use allthe N_(CM) orthogonal codes. In other words, a coded Doppler multiplexedsignal for which the number of coded Doppler multiplexes is set to besmaller than N_(CM) uses some of the orthogonal codes (for example, oneorthogonal code). Thus, code separation signal power sumPow_DeMUL^(ncm)(f_(b_cfar), f_(s_comp_cfar)±(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD) corresponding to a coded Doppler multiplexed signal(Doppler shift amount: DOP_(ndm1)) for which the number of coded Dopplermultiplexes is set to be smaller than N_(CM), where ncm=1, . . . ,N_(CM) and DopCase=1, . . . , N_(DM), contains a component whosereceived power has as low power value as about the noise level.

As an example, a description will be given of a case where the number ofcoded Doppler multiplexes of only a coded Doppler multiplexed signalthat uses DOP_(ndm1) is set to (N_(CM)−1), which is smaller than N_(CM)(see, for example, the examples of the assignment of Doppler shiftamounts and orthogonal codes illustrated in FIGS. 3 and 4 ).

In this case, coded Doppler demultiplexer 212 detects, amongPow_DeMUL^(ncm)(f_(b_cfar), f_(s_comp_cfar)±(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD), where ncm=1, . . . , N_(CM) and DopCase=1, . . . ,N_(DM), Pow_DeMUL^(ncm_min)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase_min)−1)×ΔFD) with theminimum received power. Here, “ncm_min” and “DopCase_min” represent theindex numbers of ncm and DopCase for which Pow_DeMUL^(ncm)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) has the minimumreceived power.

As another example, a description will be given of a case where thenumber of coded Doppler multiplexes of only a coded Doppler multiplexedsignal that uses DOP_(ndm1) is set to ((N_(CM)−2), which is smaller thanN_(CM) (see, for example, the examples of the assignment of Dopplershift amounts and orthogonal codes illustrated in FIGS. 7A, 7B, 8A, and8B).

In this case, among Pow_DeMUL^(ncm)(f_(b_cfar),f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD), where ncm=1, . .. , N_(CM) and DopCase=1, . . . , N_(DM), a certain DopCase does not usetwo codes (for example, the orthogonal codes indicated by crosses (“x”)in FIGS. 7A, 7B, 8A, and 8B)) among the N_(CM) orthogonal codes and thuscontains two components whose received power has as low power value asabout the noise level.

Accordingly, coded Doppler demultiplexer 212 detectsPow_DeMUL^(ncm_min1)(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase_min)−1)×ΔFD)+Pow_DeMUL^(ncm_min2)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) for which, for acombination of two different orthogonal codes Code_(ncm1) andCode_(ncm2) (where ncm1≠ncm2) among the N_(CM) orthogonal codes, powersum Pow_DeMUL^(ncm1)(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD)+Pow_DeMUL^(ncm2)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) has the minimumreceived power. Here, “ncm_min1”, “ncm_min2”, and “DopCase_min”represent the index numbers of ncm1, ncm2, and DopCase for whichPow_DeMUL^(ncm1) (f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD)+Pow_DeMUL^(ncm2)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) has the minimumreceived power.

Also when the number of coded Doppler multiplexes is set to be smallerthan (N_(CM)−2), coded Doppler demultiplexer 212 detects a combinationof candidate codes with the minimum code separation signal power sumamong combinations of candidate codes that are not used among the N_(CM)orthogonal codes. Accordingly, a combination of DopCase_min and codes(ncm_min1, ncm_min2, . . . ) can be detected.

Coded Doppler demultiplexer 212 determines, based on the detectionresult described above, that a correspondence pattern of N_(DM) codedDoppler multiplexed signals and the outputs of the respective Doppleranalyzers 210 for Doppler frequency indices (f_(s_comp_cfar)),(f_(s_comp_cfar)+ΔFD), (f_(s_comp_cfar)+2ΔFD), . . . , and(f_(s_comp_cfar)±(N_(DM)−1)ΔFD) of the N_(DM) coded Doppler multiplexedsignals for distance index f_(b_cfar) output from CFAR section 211 isDopCase_min among DopCase=1 to N_(DM).

Further, coded Doppler demultiplexer 212 determines that a coded Dopplermultiplexed signal transmitted using DOP_(ndm1) for which the number ofcoded Doppler multiplexes is set to be smaller than N_(CM) in thepattern corresponding to DopCase_min is the output of Doppler analyzer210 for distance index f_(b_cfar) and Doppler frequency index(f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase_min)−1)×ΔFD).

A reduction in the SNR of a reception signal that is a reflected wavemay make it difficult to discriminate between a signal power level and anoise power level. To address this difficulty, coded Dopplerdemultiplexer 212 may introduce a determination condition and perform aprocess of, for example, adopting a determination result (in otherwords, detection result) when the determination condition is satisfiedand removing (in other words, not adopting) a determination result whenthe determination condition is not satisfied. This can reduce theprobability of erroneous detection of the noise component and the like.For example, coded Doppler demultiplexer 212 may adopt, as adetermination result, detection value P_(MIN) with the minimum receivedpower when satisfying PowerFT(f_(b_cfar),f_(s_comp_cfa))>LEV_(DETECT)×P_(MIN). Here, LEV_(DETECT) is adetermination threshold. LEV_(DETECT) is a real number satisfying0<LEV_(DE0TECT)<1.

<(2) Process for Discrimination of Transmit Antennas 108 andDiscrimination of Doppler Frequencies of Target>

For example, coded Doppler demultiplexer 212 detects Doppler frequencyf_(TARGET) of the target in the range of −1/(2×Tr)≤f_(TARGET)<1/(2×Tr).

For example, coded Doppler demultiplexer 212 determines the presence orabsence of aliasing as in Case (a) and Case (b) below, based on thedetermination result of DopCase (for example, DopCase_min) and thedetermination result of the code whose received power is minimum (forexample, ncm_min). Then, coded Doppler demultiplexer 212 discriminatestransmit antennas 108 and Doppler frequencies of the target.

A plurality of codes whose received power is minimum (for example,ncm_min1, ncm_min2, . . . ) are also applicable by replacing ncm_min inthe following description with (ncm_min1, ncm_min2, . . . ).

[Case (a): Case without Aliasing]

For example, when ncm_min matches the index of an orthogonal codesequence that is not used (not assigned) for coded Doppler multiplexingin DOP_(ndm1) for which the number of coded Doppler multiplexes is setto be smaller than N_(CM), coded Doppler demultiplexer 212 determinesthat no Doppler aliasing occurs. In other words, when a coded Dopplermultiplexed signal corresponding to DOP_(ndm1) is not coded usingorthogonal code sequence Code_(ncm_min), coded Doppler demultiplexer 212determines that no Doppler aliasing occurs.

If it is determined that no Doppler aliasing occurs, Coded Dopplerdemultiplexer 212 determines the Doppler frequency of the target and atransmit antenna in the following way.

Doppler Frequency Determination for Target:

Coded Doppler demultiplexer 212 determines that the Doppler frequencyindex of the target is (f_(s_comp_cfar)(DopCase_min−1)×ΔFD−DOP₁). Forexample, the Doppler frequency interval for Doppler frequency indexf_(s_comp_cfar) is 1/(Ncode×Loc×Tr). Therefore, coded Dopplerdemultiplexer 212 determines that Doppler frequency f_(TARGET) of thetarget is (f_(s_comp_cfar) (DopCase_min−1)×ΔFD−DOP₁)/(Ncode×Loc×Tr).

Transmit Antenna Determination:

Coded Doppler demultiplexer 212 determines that for the output ofDoppler analyzer 210 for distance index f_(b_cfar) and Doppler frequencyindex (f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD) in thez-th signal processor 206, code separation signal DeMUL_(z)^(ncm)(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm),DopCase_min)−1)×ΔFD) using Code_(ncm) is the reception signalcorresponding to transmit antenna Tx #[ncm, ndm].

[Case (b): Case with Aliasing]

For example, when ncm_min does not match the index of an orthogonal codesequence that is not used for coded Doppler multiplexing in DOP_(ndm1)for which the number of coded Doppler multiplexes is set to be smallerthan N_(CM), coded Doppler demultiplexer 212 determines that Doppleraliasing occurs. In other words, when a coded Doppler multiplexed signalcorresponding to DOP_(ndm1) includes a multiplexed signal coded usingorthogonal code sequence Code_(ncm_min), coded Doppler demultiplexer 212determines that the output of Doppler analyzer 210 for distance indexf_(b_cfar), Doppler frequency index (f_(s_comp_cfar)±(DOPposi(DOP_(ndm),DopCase_min)−1)×ΔFD) in the z-th signal processor 206 exceeds the rangeof maximum Doppler frequencies without causing aliasing derived from thesampling theorem.

If it is determined that Doppler aliasing occurs, coded Dopplerdemultiplexer 212 decides (in other words, fixes) a Doppler frequencyindex of the target and a transmit antenna.

Doppler Frequency Determination for Target:

Coded Doppler demultiplexer 212 determines that the Doppler frequencyindex of the target is(f_(s_comp_cfar)(DopCase_min−1)×ΔFD)−DOP₁−Ncode×Sign(f_(s_comp_cfar)+(DopCase_min−1)×ΔFD−DOP₁)).For example, the Doppler frequency interval for Doppler frequency indexf_(s_comp_cfar) is 1/(Ncode×Loc×Tr). Therefore, coded Dopplerdemultiplexer 212 determines that Doppler frequency f_(TARGET) of thetarget is(f_(s_comp_cfar)+(DopCase_min−1)×ΔFD−DOP₁−Ncode×Sign(f_(s_comp_cfar)+(DopCase_min−1)×ΔFD−DOP₁))/(Ncode×Loc×Tr).Sign(x) is a sign function and is a function for real numbers x,providing as output 1 when x>0, 0 when x=0, and −1 when x<0.

In this manner, when the outputs of Doppler analyzers 210 containDoppler aliasing, coded Doppler demultiplexer 212 determines a Dopplerfrequency of the target, with consideration given to the aliasingcomponent (for example,Ncode×Sign(f_(s_comp_cfar)+(DopCase_min−1)×ΔFD−DOP₁)).

Transmit Antenna Determination:

When the outputs of Doppler analyzers 210 contain Doppler aliasing,incorrect phase correction is performed by phase correction used in thecode separation process (for example, the exp term in equation 35). Thisequivalently indicates that a code separation process is performed in acase where the elements of an orthogonal code sequence used for the codeseparation process are code elements given by following equations 37 and38.

$\begin{matrix}\lbrack 37\rbrack &  \\\{ {{O{C_{ncm}(1)}},{O{C_{ncm}(2)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{2 - 1}{Loc}} \rbrack}},\ldots,\text{ }{O{C_{ncm}( {Loc} )}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{{Loc} - 1}{Loc}} \rbrack}}} \} & ( {{Equation}37} )\end{matrix}$ $\begin{matrix}\lbrack 38\rbrack &  \\{f_{est} = {f_{{s\_ comp}{\_ cfar}} + {( {{DopCase\_ min} - 1} ) \times \Delta FD} - {DOP_{1}}}} & ( {{Equation}38} )\end{matrix}$

For example, in a case where N_(CM)=2 and orthogonal code sequenceCode₁={1, 1} with code length Loc=2 is used, a process equivalent to aseparation process using Code₂={1, −1}, as given by following equation39, is performed, regardless of Sign(f_(est)).

$\begin{matrix}\lbrack 39\rbrack &  \\{\{ {{O{C_{1}(1)}},{O{C_{1}(2)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{1}{2}} \rbrack}}} \} = {\{ {1,{1 \times {\exp\lbrack {{- j}\pi{Sign}( f_{est} )} \rbrack}}} \} = {\{ {1,{- 1}} \} = {Code}_{2}}}} & ( {{Equation}39} )\end{matrix}$

In a case where Code₂={1, −1} is used, on the other hand, a processequivalent to a separation process using Code₁={1, 1}, as given byfollowing equation 40, is performed, regardless of Sign(f_(est)).

$\begin{matrix}\lbrack 40\rbrack &  \\{\{ {{O{C_{2}(1)}},{O{C_{2}(2)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{1}{2}} \rbrack}}} \} = {\{ {1,{{- 1} \times {\exp\lbrack {j\pi{Sign}( f_{est} )} \rbrack}}} \} = {\{ {1,1} \} = {Code}_{1}}}} & ( {{Equation}40} )\end{matrix}$

Accordingly, in a case where N_(CM)=2 and orthogonal code sequencesCode₁={1, 1} and Code₂={1, −1} with code length Loc=2 are used, codedDoppler demultiplexer 212 determines that for the output of Doppleranalyzer 210 for distance index f_(b_cfar) and Doppler frequency index(f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD) in the z-thsignal processor 206, code separation signal DeMUL_(z) ¹(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD) using Code₁ isthe reception signal of the reflected wave transmitted from transmitantenna Tx #[2, ndm] and code separation signal DeMUL_(z) ²(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD) using Code₂ isthe reception signal of the reflected wave transmitted from transmitantenna Tx #[1, ndm].

For example, in a case where N_(CM)=3 and orthogonal code sequencesCode₁={1, 1, 1}, Code₂={1, exp(j2π/3), exp(j4π/3)}, and Code₃={1,exp(−j2π/3), exp(−j4π/3)} with code length Loc=3 (=N_(CM)) are used,determination is made as following.

For example, in a case where Code₁={1, 1, 1} is used, as given byfollowing equation 41, a process equivalent to a separation processusing Code₃ or Code₂ in accordance with positive or negativeSign(f_(est)) is performed.

$\begin{matrix}\lbrack 41\rbrack &  \\{\{ {{O{C_{1}(1)}},{O{C_{1}(2)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{1}{3}} \rbrack}},{O{C_{1}(3)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{2}{3}} \rbrack}}} \} = \{ \begin{matrix}{{Code}_{3},{{{if}{Sign}( f_{est} )} > 0}} \\{{Code}_{2},{{{if}{Sign}( f_{est} )} < 0}}\end{matrix} } & ( {{Equation}41} )\end{matrix}$

For example, in a case where Code₂={1, exp(j2π/3), exp(j4π/3)} is used,as given by following equation 42, a process equivalent to a separationprocess using Code₁ or Code₃ in accordance with positive or negativeSign(f_(est)) is performed.

$\begin{matrix}\lbrack 42\rbrack &  \\{\{ {{O{C_{2}(1)}},{O{C_{2}(2)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{1}{3}} \rbrack}},{O{C_{2}(3)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{2}{3}} \rbrack}}} \} = \{ \begin{matrix}{{Code}_{1},{{{if}{Sign}( f_{est} )} > 0}} \\{{Code}_{3},{{{if}{Sign}( f_{est} )} < 0}}\end{matrix} } & ( {{Equation}42} )\end{matrix}$

For example, in a case where Code₃={1, exp(−j2π/3), exp(−j4π/3)} isused, as given by following equation 43, a process equivalent to aseparation process using Code₂ or Code₁ in accordance with positive ornegative Sign(f_(est)) is performed.

$\begin{matrix}\lbrack 43\rbrack &  \\{\{ {{O{C_{3}(1)}},{O{C_{3}(2)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{1}{3}} \rbrack}},{O{C_{3}(3)}{\exp\lbrack {{- j}2\pi{Sign}( f_{est} )\frac{2}{3}} \rbrack}}} \} = \{ \begin{matrix}{{Code}_{2},{{{if}{Sign}( f_{est} )} > 0}} \\{{Code}_{1},{{{if}{Sign}( f_{est} )} < 0}}\end{matrix} } & ( {{Equation}43} )\end{matrix}$

Accordingly, in a case where N_(CM)=3 and orthogonal code sequencesCode₁={1, 1, 1}, Code₂={1, exp(j2π/3), exp(j4π/3)}, and Code₃={1,exp(−j2π/3), exp(−j4π/3)} with code length Loc=3 (=N_(CM)) are used,coded Doppler demultiplexer 212 determines transmit antenna 108 in thefollowing way for the output of Doppler analyzer 210 for distance indexf_(b_cfar) and Doppler frequency index(f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD) in the z-thsignal processor 206.

Case where Sign(f_(est))>0:

Coded Doppler demultiplexer 212 determines that code separation signalDeMUL_(z) ¹(f_(b_cfar), f_(s_comp_cfar)±(DOPposi(DOP_(ndm),DopCase_min)−1)×ΔFD) using Code₁ is the reception signal of thereflected wave transmitted from transmit antenna Tx #[3, ndm], that codeseparation signal DeMUL_(z) ²(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD) using Code₂ isthe reception signal of the reflected wave transmitted from transmitantenna Tx #[1, ndm], and that code separation signal DeMUL_(z)³(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD)using Code₃ is the reception signal of the reflected wave transmittedfrom transmit antenna Tx #[2, ndm].

Case where Sign(f_(est))<0:

Coded Doppler demultiplexer 212 determines that code separation signalDeMUL_(z) ¹(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm),DopCase_min)−1)×ΔFD) using Code₁ is the reception signal of thereflected wave transmitted from transmit antenna Tx #[2, ndm], that codeseparation signal DeMUL_(z) ²(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD) using Code₂ isthe reception signal of the reflected wave transmitted from transmitantenna Tx #[3, ndm], and that code separation signal DeMUL_(z)³(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD)using Code₃ is the reception signal of the reflected wave transmittedfrom transmit antenna Tx #[1, ndm].

As described above, the correspondence between a code used for codingduring transmission and a code separated by the code separation processduring aliasing determination is decided in advance based on orthogonalcode sequence Code_(ncm) (ncm=1, . . . , N_(CM)) and Sign(f_(est)). Thisenables coded Doppler demultiplexer 212 to use, for example, codeconversion function AliasConv[ncm, Sign(f_(est))]=Tx_ncm (that outputsorthogonal code sequence Code_(Tx_ncm) index Tx_ncm separated in thecode separation process using Code_(ncm) during aliasing determination).

Further, inverse code conversion function AliasConv⁻¹[Tx_ncm,Sign(f_(est))]=ncm (that outputs index ncm of code sequence Code_(ncm)used in the code separation process, where the index of an orthogonalcode sequence separated in the code separation process during aliasingdetermination is Tx_ncm) can also be defined in a similar manner.

For example, in a case where N_(CM)=2 and orthogonal code sequencesCode₁={1, 1} and Code₂={1, −1} with code length Loc=2 are used, codeconversion function AliasConv[ncm, Sign(f_(est))] is decided in advancesuch that AliasConv[1, Sign(f_(est))]=2 and AliasConv[2,Sign(f_(est)]=1. Further, the inverse code conversion function isdecided in advance such that AliasConv⁻¹[1, Sign(f_(est))]=2 andAliasConv⁻¹[2, Sign(f_(est))]=1.

For example, in a case where N_(CM)=3 and orthogonal code sequencesCode₁={1, 1, 1}, Code₂={1, exp(j2π/3), exp(j4π/3)}, and Code₃={1,exp(−j2π/3), exp (−j4π/3)} with code length Loc=3 (=N_(CM)) are used,code conversion function AliasConv[ncm, Sign(f_(est))] is decided inadvance in the following way.

Case where Sign(f_(est))>0:

Code conversion function AliasConv[ncm, Sign(f_(est))] is decided inadvance such that AliasConv[1, Sign(f_(est))]=3, AliasConv[2,Sign(f_(est))]=1, and AliasConv[3, Sign(f_(est))]=2. Further, theinverse code conversion function is decided in advance such thatAliasConv⁻¹[1, Sign(f_(est))]=2, AliasConv⁻¹[2, Sign(f_(est))]=3, andAliasConv⁻¹[3, Sign(f_(est))]=1.

Case where Sign(f_(est))<0:

Code conversion function AliasConv[ncm, Sign(f_(est))] is decided inadvance such that AliasConv[1, Sign(f_(est))]=2, AliasConv[2,Sign(f_(est))]=3, and AliasConv[3, Sign(f_(est))]=1. Further, theinverse code conversion function is decided in advance such thatAliasConv⁻¹[1, Sign(f_(est))]=3, AliasConv⁻¹[2, Sign(f_(est))]=1, andAliasConv⁻¹[3, Sign(f_(est))]=2.

Accordingly, for example, with the use of code conversion functionAliasConv[ncm, Sign(f_(est))], coded Doppler demultiplexer 212 performstransmit antenna determination during aliasing determination in thefollowing way. For example, coded Doppler demultiplexer 212 determinesthat for the output of Doppler analyzer 210 for distance indexf_(b_cfar) and Doppler frequency index(f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_min)−1)×ΔFD) in the z-thsignal processor 206, code separation signal DeMUL_(z)^(ncm)(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm),DopCase_min)−1)×ΔFD) using Code_(ncm) is the reception signalcorresponding to transmit antenna Tx #[AliasConv[ncm, Sign(f_(est))],ndm].

Alternatively, coded Doppler demultiplexer 212 may determine that thereception signal corresponding to transmit antenna Tx #[Tx_ncm, ndm] isa code separation signal given by following equation 44 that usesCode_(ncm) (where ncm=AliasConv⁻¹[Tx_ncm, Sign(f_(est))]).

[44]

DeMUL_(z) ^(AliasConv) ⁻¹ ^([Tx_ncm,Sign(f) ^(est) ^()])(f _(b_cfar) ,f_(s_comp_cfar)+(DOPposi(DOP_(ndm),DopCase_min)−1)×ΔFD)   (Equation 44)

Also in a case where N_(CM)>3, the code conversion function and theinverse code conversion function can be determined in advance, and thedetermination of a transmit antenna can be performed in a similarmanner.

In this embodiment, accordingly, radar receiver 200 can perform aliasingdetermination (for example, determination of the presence or absence ofaliasing and determination of correspondence DopCase) based on theresult of code separation of a coded Doppler multiplexed signal forwhich the number of coded Doppler multiplexes is set to be smaller thanN_(CM). This enables radar apparatus 10 to discriminate transmitantennas 108 corresponding to coded Doppler multiplexed signals and todiscriminate Doppler frequencies of the target even if aliasing occurs.According to this embodiment, therefore, for example, the range overwhich a Doppler frequency is detectable without ambiguity can beextended to the range greater than or equal to −1/(2Tr) and less than1/(2Tr).

For example, the range over which a Doppler frequency is detectablewithout ambiguity when one transmit antenna 108 is used for transmissionis the range greater than or equal to −1/(2Tr) and less than 1/(2Tr). Inthis embodiment, therefore, even when plural transmit antennas 108 areused, the range over which a Doppler frequency is detectable withoutambiguity can be achieved in a manner similar to that when a singleantenna is used for transmission.

Next, processing for a plurality of coded Doppler multiplexed signalsfor which the numbers of coded Doppler multiplexes are set to be smallerthan N_(CM) (see, for example, FIGS. 5A to 5C, 6A to 6C, 10A, 10B, 11A,11B, and so on) will be described.

For example, a coded Doppler multiplexed signal for which the number ofcoded Doppler multiplexes is set to be smaller than N_(CM) is not onlyDoppler shift amount DOP_(nd) but there is a plurality of coded Dopplermultiplexed signals for which the numbers of coded Doppler multiplexesare set to be smaller than N_(CM). In this case, coded Dopplerdemultiplexer 212 detects ncm_min1, ncm_min2, and DopCase_min in thefollowing way.

For example, when coded Doppler multiplexed signals for which thenumbers of coded Doppler multiplexes are set to (N_(CM)−1), which issmaller than N_(CM), are Doppler shift amounts DOP_(ndm1) andDOP_(ndm2), coded DOP Doppler demultiplexer 212 calculates, based onequation 36, code separation signal power sumsPow_DeMUL^(ncm)(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD) and Pow_DeMUL^(ncm)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm2), DopCase)−1)×ΔFD). Here, ncm=1, . .. , N_(CM), and DopCase=1, . . . , N_(DM).

Here, in a coded Doppler multiplexed signal that uses Doppler shiftamount DOP_(ndm1), an orthogonal code that is not used for coded Dopplermultiplexing is represented by Code_(ncm1), and in a coded Dopplermultiplexed signal that uses Doppler shift amount DOP_(ndm2), anorthogonal code that is not used for coded Doppler multiplexing isrepresented by Code_(ncm2).

In this case, in code separation signal power sumPow_DeMUL^(ncm1)(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD)+Pow_DeMUL^(ncm2)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm2), DopCase)−1)×ΔFD) or code separationsignal power sum Pow_DeMUL^(AliasConv[ncm1, Sign(fest)])(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD)+Pow_DeMUL^(AliasConv[ncm2, Sign(fest)])(f_(b_cfar),f_(s_comp_cfar)±(DOPposi(DOP_(ndm2), DopCase)−1)×ΔFD) during Doppleraliasing determination, coded Doppler demultiplexer 212 detectsPow_DeMUL^(ncm_min1)(f_(b_cfar), f_(s_comp_cfar)±(DOPposi(DOP_(ndm1),DopCase_min)−1)×ΔFD)+Pow_DeMUL^(ncm_min2)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm2), DopCase_min)−1)×ΔFD) with theminimum received power when DopCase is equal to any of 1, . . . , andN_(DM).

Here, “ncm_min1”, “ncm_min2”, and “DopCase_min” represent the indexnumbers of ncm1, ncm2, and DopCase for which code separation signalpower sum Pow_DeMUL^(ncm1)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD)+Pow_DeMUL^(ncm2)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm2), DopCase)−1)×ΔFD) has the minimumreceived power.

The number of coded Doppler multiplexed signals for which the numbers ofcoded Doppler multiplexes are set to be smaller than N_(CM) is notlimited to two and may be three or more.

The foregoing description has been given of an example of the operationof coded Doppler demultiplexer 212.

In FIG. 1 , direction estimator 213 performs a direction estimationprocess for the target based on distance index f_(b_cfar) input fromcoded Doppler demultiplexer 212, the output of Doppler analyzer 210 forDoppler frequency index (f_(s_comp_cfar)±(DOPposi(DOP_(ndm),DopCase_min)−1)×ΔFD), the Doppler frequency determination result for thetarget, and the transmit antenna determination result (or determinationresult of Doppler aliasing).

For example, direction estimator 213 generates, based on the output ofcoded Doppler demultiplexer 212, virtual receive array correlationvector h(f_(b_cfar), f_(s_comp_cfar)) given by following equation 45 or46 and performs a direction estimation process.

Virtual receive array correlation vector h(f_(b_cfar), f_(s_comp_cfar))includes Nt×Na elements, the number of which is equal to the product ofthe number Nt of transmit antennas and the number Na of receiveantennas. Virtual receive array correlation vector h(f_(b_cfar),f_(s_comp_cfar)) is used in a process for performing directionestimation on reflected wave signals from the target, the directionestimation being based on a phase difference among receive antennas 202.Here, z=1, . . . , Na.

Case without Aliasing:

$\begin{matrix}\lbrack 45\rbrack &  \\{{h( {f_{b\_ cfar},f_{{s\_ comp}{\_ cfar}}} )} = \begin{bmatrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}{{DeMUL}_{1}^{1}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )} \\{{DeMUL}_{2}^{1}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \end{matrix} \\{{DeMUL}_{Na}^{1}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \end{matrix} \\{{DeMUL}_{1}^{N_{{{DOP}\_{CODE}}^{(1)}}}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\{{DeMUL}_{2}^{N_{{{DOP}\_{CODE}}^{(1)}}}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \end{matrix} \\{{DeMUL}_{Na}^{N_{{{DOP}\_{CODE}}^{(1)}}}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \end{matrix} \\{{DeMUL}_{1}^{1}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\{{DeMUL}_{2}^{1}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \end{matrix} \\{{DeMUL}_{Na}^{1}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \end{matrix} \\{{DeMUL}_{1}^{N_{{{DOP}\_{CODE}}^{(N_{DM})}}}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\{{DeMUL}_{2}^{N_{{{DOP}\_{CODE}}^{(N_{DM})}}}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \end{matrix} \\{{DeMUL}_{Na}^{N_{{{DOP}\_{CODE}}^{(N_{DM})}}}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{bmatrix}} & ( {{Equation}45} )\end{matrix}$

Case with Aliasing:

$\begin{matrix}\lbrack 46\rbrack &  \\{{h( {f_{b\_ cfar},f_{{s\_ comp}{\_ cfar}}} )} = \begin{bmatrix}\begin{matrix}\begin{matrix}\begin{matrix}{{DeMUL}_{1}^{{AliasConv}^{- 1}\lbrack{1,{{Sign}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )} \\{{DeMUL}_{2}^{{AliasConv}^{- 1}\lbrack{1,{{Sign}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \end{matrix} \\{{DeMUL}_{Na}^{{AliasConv}^{- 1}\lbrack{1,{{Sign}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )}\end{matrix} \\ \vdots \\\begin{matrix}{{DeMUL}_{1}^{{AliasConv}^{- 1}\lbrack{{N_{DOP\_ CODE}(1)},{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} +}} } \\ {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}} ) \\\begin{matrix}{{DeMUL}_{2}^{{AliasConv}^{- 1}\lbrack{{N_{DOP\_ CODE}(1)},{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} +}} } \\ {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}} ) \\ \vdots \\\begin{matrix}{{DeMUL}_{Na}^{{AliasConv}^{- 1}\lbrack{{N_{DOP\_ CODE}(1)},{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} +}} } \\ {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}} ) \\\begin{matrix} \vdots \\{{DeMUL}_{1}^{{AliasConv}^{- 1}\lbrack{1,{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )} \\{{DeMUL}_{2}^{{AliasConv}^{- 1}\lbrack{1,{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{1},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} )} \\ \vdots \\{{DeMUL}_{Na}^{{AliasConv}^{- 1}\lbrack{1,{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}}}} } \\ \vdots \\\begin{matrix}{{DeMUL}_{1}^{{AliasConv}^{- 1}\lbrack{{N_{DOP\_ CODE}(N_{DM})},{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} +}} } \\ {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}} ) \\\begin{matrix}{{DeMUL}_{2}^{{AliasConv}^{- 1}\lbrack{{N_{DOP\_ CODE}(N_{DM})},{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} +}} } \\ {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}} ) \\ \vdots \\\begin{matrix}{{DeMUL}_{Na}^{{AliasConv}^{- 1}\lbrack{{N_{DOP\_ CODE}(N_{DM})},{{S{ign}}(f_{est})}}\rbrack}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} +}} } \\ {\{ {{{DOPposi}( {{DOP}_{N_{DM}},{DopCase\_ min}} )} - 1} \} \times \Delta{FD}} )\end{matrix}\end{matrix}\end{matrix}\end{matrix}\end{matrix}\end{matrix}\end{matrix}\end{bmatrix}} & ( {{Equation}46} )\end{matrix}$

For example, direction estimator 213 calculates a spatial profile, withazimuth direction θ in direction estimation evaluation function valueP_(H)(θ, f_(b_cfar), f_(s_comp_cfar)) being variable within a definedangular range. Direction estimator 213 extracts a predetermined numberof local maximum peaks in the calculated spatial profile in order fromthe largest and outputs the azimuth directions of the local maximumpeaks as direction-of-arrival estimation values (for example, positionmeasurement output).

There are various methods with direction estimation evaluation functionvalue P_(H)(θ, f_(b_cfar), f_(s_comp_cfar)) depending on thedirection-of-arrival estimation algorithm. For example, an estimationmethod using an array antenna, as disclosed in NPL 3, may be used.

For example, when a virtual receive array includes Nt×Na elementsarranged in a straight line at equal intervals d_(H), a beamformermethod can be given by following equations 47 and 48. Any othertechnique, such as Capon or MUSIC, may be applicable.

$\begin{matrix}\lbrack 47\rbrack &  \\{{P_{H}( {\theta_{u},f_{b\_ cfar},f_{{s\_ comp}{\_ cfar}}} )} = {❘{{a^{H}( \theta_{u} )}D_{cal}{h( {f_{b\_ cfar},f_{{s\_ comp}{\_ cfar}}} }}❘}^{2}} & ( {{Equation}47} )\end{matrix}$ $\begin{matrix}\lbrack 48\rbrack &  \\{{a( \theta_{u} )} = \begin{bmatrix}\begin{matrix}\begin{matrix}1 \\{\exp\{ {{- j}2\pi d_{H}\sin\theta_{u}/\lambda} \}}\end{matrix} \\ \vdots \end{matrix} \\{\exp\{ {{- j}2\pi( {{N_{t}N_{a}} - 1} )d_{H}\sin\theta_{u}/\lambda} \}}\end{bmatrix}} & ( {{Equation}48} )\end{matrix}$

In equation 47, the superscript H denotes the Hermitian transposeoperator. Further, a(θ_(u)) denotes the direction vector of the virtualreceive array relative to an incoming wave in azimuth direction θ_(u).

Azimuth direction θ_(u) is a vector that is changed at azimuth intervalβ₁ in an azimuth range over which direction-of-arrival estimation isperformed. For example, θ_(u) is set as follows.

θ_(u)=θmin+uβ ₁ , u=0, . . . ,NU

NU=floor[(θmax−θmin)/β₁]+1

Here, floor(x) is a function that returns the largest integer value notgreater than real number x.

In equation 47, furthermore, Dcal is an (Nt×Na) order matrix includingan array correction coefficient for correcting phase deviations andamplitude deviations across the transmit array antenna and across thereceive array antenna, and a coefficient for reducing the influence ofcoupling of elements across the antennas. If the coupling betweenantennas in the virtual receive array is negligible, Dcal is a diagonalmatrix with diagonal components including an array correctioncoefficient for correcting phase deviations and amplitude deviationsacross the transmit array antenna and across the receive array antenna.

For example, direction estimator 213 may output, as position measurementresults, distance information based on distance index f_(b_cfar) andDoppler velocity information of the target based on the Dopplerfrequency determination result for the target, together with thedirection estimation result, to, for example, a vehicle controlapparatus (not illustrated) in the case of an in-vehicle radar or to aninfrastructure control apparatus (not illustrated) in the case of aninfrastructure radar.

Doppler frequency information may be converted into the relativevelocity component and then output. Doppler frequency index f_(our) inthe Doppler frequency determination result for the target can beconverted into relative velocity component v_(d)(f_(out)) usingfollowing equation 49. Here, λ denotes the wavelength of the carrierfrequency of an RF signal output from a transmission radio section (notillustrated). Further, Δ_(f) denotes the Doppler frequency interval inFFT processing performed in Doppler analyzer 210. For example, in thisembodiment, Δ_(f)=1/{N_(code)×Loc×T_(r)}.

$\begin{matrix}\lbrack 49\rbrack &  \\{{v_{d}( f_{out} )} = {\frac{\lambda}{2}f_{out}\Delta_{f}}} & ( {{Equation}49} )\end{matrix}$

As described above, in this embodiment, radar apparatus 10 applies phaserotation amounts corresponding to Doppler shift amounts and orthogonalcode sequences to radar transmission signals to transmit radartransmission signals (in other words, coded Doppler multiplexed signals)from plural transmit antennas 108 in a multiplexed manner. In thisembodiment, plural transmit antennas 108 are associated withcombinations of Doppler shift amounts (DOP_(ndm)) and orthogonal codesequences (DOP_(ncm)) such that in each of the combinations, at leastone of Doppler shift amount (DOP_(ndm)) or orthogonal code sequence(DOP_(ncm)) is different. In this embodiment, furthermore, the number ofmultiplexes of the orthogonal code sequence (in other words, the numberof codes) corresponding to each Doppler shift amount in combinations ofDoppler shift amounts and orthogonal code sequences is different. Inother words, the numbers of coded Doppler multiplexes for the respectiveDoppler-multiplexed transmission signals are set to be non-uniform.

Radar apparatus 10 can determine, based on, for example, the receivedpower of a code-separated signal for each coded Doppler multiplexedsignal, transmit antenna 108 associated with the coded Dopplermultiplexed signal (in other words, the combination of Doppler shiftamount and orthogonal code sequence) and the presence or absence ofDoppler aliasing (for example, DopCase and the like). This enables radarapparatus 10 to appropriately determine a Doppler frequency of thetarget even in the presence of Doppler aliasing.

According to this embodiment, therefore, radar apparatus 10 can extendthe effective Doppler frequency bandwidth to 1/(Tr) and can extend theDoppler frequency (relative velocity) detection range with no ambiguity.Accordingly, radar apparatus 10 can improve target-object sensingaccuracy over a wider Doppler frequency range.

In this embodiment, furthermore, coded Doppler multiplexing, which isperformed using both Doppler multiplexing and coding, can reduce thenumber of Doppler multiplexes compared with the use of only Dopplermultiplexing in multiplexing transmission. This can increase theintervals of phase rotation amounts for applying Doppler shifts,thereby, for example, relieving the accuracy requirements (phasemodulation accuracy) for the phase shifters and achieving the costreduction effect of an RF section, including reduction of the man-hoursrequired for the adjustment of the phase shifters.

In this embodiment, furthermore, since coded Doppler multiplexing isperformed using both Doppler multiplexing and coding, radar apparatus 10performs, for each code element, Fourier frequency analysis (FFTprocessing) to detect the Doppler frequency (detect the relativevelocity). Accordingly, for example, compared with Fourier frequencyanalysis (FFT processing) to detect the Doppler frequency (detect therelative velocity) using only Doppler multiplexing in multiplexingtransmission, the FFT size corresponds to (1/code length) and the numberof times FFT processing is performed is increased by (code length)times. For example, when the FFT operation amount with FFT size Nc isroughly estimated to be Nc×log₂(Nc), coded Doppler multiplexingaccording to this embodiment has an operation amount ratio of about{Loc×Nc/Loc×log₂(Nc/Loc)}/{Nc×log₂(Nc)}=1−log₂(Loc)/log₂(Nc) relative toFFT operation with only Doppler multiplexing. For example, in a casewhere Loc=2 and Nc=1024, the operation amount ratio is 0.9. Theoperation reduction effect of FFT processing can be achieved, and theeffect of simplification of the circuit configuration and cost reductioncan also be achieved.

(Variation 1 of Embodiment 1)

Phase rotation amount ϕ_(ndm) for applying Doppler shift amountDOP_(ndm) is not limited to, for example, the value given in equation 1and so on. For example, phase rotation amount ϕ_(ndm) may be a valuegiven by following equation 50. Here, round(x) represents the roundfunction that outputs a rounded integer value for real number x. Theround(N_(code)/N_(DM)) term is introduced so that the phase rotationamount is an integer multiple of the Doppler frequency interval inDoppler analyzer 210. In equation 50, the angle is expressed in radian.

$\begin{matrix}\lbrack 50\rbrack &  \\{\phi_{ndm} = {\frac{2\pi}{N_{code}}{round}( \frac{N_{code}}{N_{DM}} )( {{ndm} - 1} )}} & ( {{Equation}50} )\end{matrix}$

(Variation 2 of Embodiment 1)

Embodiment 1 has described a case where in coder 106 in phase rotationamount setter 104, the numbers of coded Doppler multiplexesN_(DOP_CODE)(1), N_(DOP_CODE)(2), . . . , and N_(DOP_CODE)(N_(DM)) areset to be different (in other words, non-uniform) in a range greaterthan or equal to 1 and less than or equal to N_(CM); however, this isnot required. For example, at least one of the numbers of coded Dopplermultiplexes N_(DOP_CODE)(1), N_(DOP_CODE)(2), . . . , andN_(DOP_CODE)(N_(DM)) may be greater than or equal to 1 and less thanN_(CM), and the number of coded Doppler multiplexes, whose value is 0,may be included.

For example, coded Doppler demultiplexer 212 detects, for combinationsof candidate codes that are not used among N_(CM) orthogonal codes, acombination with the minimum code separation signal power sum.Accordingly, combinations of DopCase_min and codes (ncm_min1, ncm_min2,. . . ) can be detected. Further, coded Doppler demultiplexer 212performs Doppler aliasing determination by utilizing the inclusion of atleast one DOP_(ndm1) for which the number of coded Doppler multiplexesis set to be greater than or equal to 1 and less than N_(CM).

This enables radar apparatus 10 to detect Doppler frequency f_(TARGET)of the target in the range of −1/(2×Tr)≤f_(TARGET)<1/(2×Tr) and toperform discrimination of transmit antennas 108 and discrimination ofDoppler frequencies of the target.

For example, FIG. 14A illustrates a case where Nt=3, N_(DM)=3, andN_(CM)=2 and where the assignment numbers (in other words, the numbersof coded Doppler multiplexes) of orthogonal codes Code₁ and Code₂ toDoppler shift amounts DOP₁, DOP₂, and DOP₃ are set such thatN_(DOP_CODE)(1)=0, N_(DOP_CODE)(2)=1, and N_(DOP_CODE)(3)=2. Also inFIG. 14A, effects similar to those in Embodiment 1 are obtained.

FIG. 14B illustrates a case where Nt=5, N_(DM)=4, and N_(CM)=2 and wherethe assignment numbers (in other words, the numbers of coded Dopplermultiplexes) of orthogonal codes Code₁ and Code₂ to Doppler shiftamounts DOP₁, DOP₂, DOP₃, and DOP₄ are set such that N_(DOP_CODE)(1)=2,N_(DOP_CODE)(2)=0, N_(DOP_CODE)(3)=2, and N_(DOP_CODE)(4)=1. Also inFIG. 14B, effects similar to those in Embodiment 1 are obtained.

(Variation 3 of Embodiment 1)

Variation 3 describes the sub-array configuration of transmit antennasof a radar apparatus.

A combination of some of the transmit antennas may be used as asub-array to narrow the beam width of a transmission directional beampattern to improve transmission directional gain. This narrows thesensible angular range, but increases the sensible distance range. Inaddition, a beam weight factor for generating a directional beam can bemade variable to control the beam direction to be variable.

FIG. 15 is a block diagram illustrating an example configuration ofradar transmitter 100 a according to Variation 3. In FIG. 15 ,components that perform operations similar to those in radar transmitter100 illustrated in FIG. 1 are identified with the same numerals, and adescription thereof is omitted. A radar receiver according to Variation3 has substantially the same basic configuration as that of radarreceiver 200 illustrated in FIG. 1 and will thus be described withreference to FIG. 1 .

In FIG. 15 , for example, sub-arrays (for example, Nt sub-arrays) eachhaving N_(SA) transmit antennas 108 are configured for the respectiveoutputs of Nt phase rotators 107. The sub-array configuration oftransmit antennas 108 is not limited to that in the example illustratedin FIG. 15 . For example, the number of transmit antennas (in otherwords, N_(SA)) included in a sub-array for the output of each phaserotator 107 may differ from phase rotator 107 to phase rotator 107.Here, N_(SA) is an integer greater than or equal to 1. In a case whereN_(SA)=1, a configuration similar to that in FIG. 1 is provided.

In FIG. 15 , beam weight generator 109 generates a beam weight fordirecting the main beam direction of transmission beams to apredetermined direction using sub-arrays. For example, the transmissionbeam direction when sub-arrays each having N_(SA) transmit antennas arearranged along a straight line at element spacing d_(SA) is representedby θ_(TxBr). In this case, for example, beam weight generator 109generates beam weight W_(Tx)(Index_TxSubArray, θ_(TxBF)) as given byfollowing equation 51.

$\begin{matrix}\lbrack 51\rbrack &  \\{{W_{Tx}( {{Index\_ TxSubArray},\theta_{TxBF}} )} = \text{ }\begin{bmatrix}\begin{matrix}\begin{matrix}1 \\{\exp( {j2\pi d\sin\theta_{TxBF}/\lambda} )}\end{matrix} \\ \vdots \end{matrix} \\{\exp\lbrack {j2\pi( {{Index\_ TxSubArray} - 1} )d_{SA}\sin\theta_{TxBF}/\lambda} \rbrack}\end{bmatrix}} & ( {{Equation}51} )\end{matrix}$

Here, Index_TxSubArray represents the element index of a sub-array, andIndex_TxSubArray=1, . . . , N_(SA). Further, λ denotes the wavelength ofa radar transmission signal, and d_(SA) denotes sub-array antennaspacing.

Each beam weight multiplier 110 multiplies the output of thecorresponding one of phase rotators 107 by beam weight factorW_(Tx)(Index_TxSubArray, θ_(TxBF)) input from beam weight generator 109.Transmission signals multiplied by beam weight W_(Tx)(Index_TxSubArray,θ_(TxBF)) are transmitted from N_(SA) sub-array antennas. Here,Index_TxSubArray=1, . . . , N_(SA).

With the operation described above, radar transmitter 100 a can performtransmission such that transmission directional beams can be directed ina predetermined direction using sub-arrays for the outputs of phaserotators 107. This can improve transmission directional gain in apredetermined direction and extend the sensible distance range.Additionally, the SNR can be improved.

Further, radar transmitter 100 a can variably set a beam weight factorfor generating a transmission directional beam to control the beamdirection to be variable.

The configuration for sub-array transmission described in Variation 3 isalso applicable to other variations or embodiments.

In the configuration illustrated in FIG. 15 , phase rotation using phaserotators 107 and beam weight multiplication using beam weightmultipliers 110 are separately performed; however, this configuration isnot required. For example, each beam weight multiplier 110 may performmultiplication such that phase rotation ψ_(ndop_code(ndm), ndm)(m)obtained by the corresponding one of phase rotators 107 is included inbeam weight factor W_(Tx)(Index_TxSubArray, θ_(TxBF)). That is, eachbeam weight multiplier 110 may multiply a chirp signal output from radartransmission signal generator 101 by W_(Tx)(Index_TxSubArray,θ_(TxBF))×exp(jψ_(ndop_code(ndm), ndm)(m)). This configuration canremove phase shifters and phase modulators of phase rotators andsimplify the circuit configuration.

(Variation 4 of Embodiment 1)

Variation 4 describes a case where the Doppler shift amount (ortransmission period Tr) varies for each transmission frame. In otherwords, the intervals of Doppler shift amounts used for coded Dopplermultiplexing transmission are each set to vary for each frame (forexample, every Nc transmission periods (Nc×Tr)) in which a radartransmission signal is transmitted.

For example, there is a plurality of targets having substantially equalreception levels for the same distance index f_(b_cfar). In this case,if the intervals of Doppler peaks of the plurality of targets match theintervals of Doppler shift amounts in Doppler multiplexing, codedDoppler demultiplexer 212 may be unable to discriminate transmitantennas 108 and the Doppler frequencies of the targets.

Since the Doppler frequencies of a plurality of targets may differ, therelative movement speeds between the targets and radar apparatus 10 maydiffer. Radar apparatus 10 continuously performs radar observation suchthat even if transmit antennas 108 and the Doppler frequencies of thetargets fail to be discriminated in radar position measurement output ata certain point in time, the distance between the plurality of targetsis different in radar position measurement output at the subsequentpoint in time, resulting in it being likely that measurement issuccessful. It is therefore assumed that separate output of each of theplurality of targets is likely to be obtained.

In addition, to more reliably separate signals corresponding to theplurality of targets, for example, radar apparatus 10 may continuouslyperform radar position measurement by setting at least one oftransmission period Tr and the Doppler shift amount to vary for eachradar position measurement (for example, every Nc transmission periods(Nc×Tr)).

Accordingly, for example, even if, for the same distance indexf_(b_cfar), the reception levels of Doppler peaks of a plurality oftargets are substantially equal, the intervals of the Doppler peaksmatch the intervals of Doppler shift amounts, and coded Dopplerdemultiplexer 212 is unable to discriminate transmit antennas 108 and todiscriminate the Doppler frequencies of the targets, the intervals ofthe Doppler shift amounts may be more likely to be different in thesubsequent radar position measurement. This enables radar apparatus 10to more reliably separate the signals corresponding to the plurality oftargets.

For example, phase rotation amount ϕ_(ndm) for applying Doppler shiftamount DOP_(ndm) may be set as given by following equation 52 (the angleis expressed in radian).

$\begin{matrix}\lbrack 52\rbrack &  \\{\phi_{ndm} = \frac{2{\pi( {{ndm} - 1} )}}{N_{DM} + \delta}} & ( {{Equation}52} )\end{matrix}$

For example, in equation 52, radar apparatus 10 varies δ for each radarposition measurement, thereby variably setting the intervals of theDoppler shift amounts. For example, radar apparatus 10 may periodicallyvary δ for each radar position measurement to 0, 1, 0, 1, . . . .

Further, for example, radar apparatus 10 sets transmission period Tr tovary for each radar position measurement, thereby changing the intervalsof the Doppler shift amounts. As a result, an effect equivalent to thatwhen the Doppler shift amounts are variably set is achieved.

(Variation 5 of Embodiment 1)

Variation 5 describes, for example, a method for reducing the influenceof interference from a plurality of radar apparatuses for which the sameor some frequency bands overlap.

FIG. 16 is a block diagram illustrating an example configuration ofradar apparatus 10 b according to Variation 5. In FIG. 16 , the samecomponents as those in FIG. 1 are identified with the same numerals, anda description thereof is omitted. For example, in radar apparatus 10 billustrated in FIG. 16 , compared with radar apparatus 10 illustrated inFIG. 1 , phase rotation amount setter 104 b in radar transmitter 100 bfurther includes random code applier 111, and signal processor 206 b inradar receiver 200 b further includes random code multiplier 214.

In FIG. 16 , random code applier 111 multiplies, for example, codedDoppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m) output fromcoder 106 by code element RC(RC_INDEX(m)) of pseudo-random code sequenceRCode and outputs the result to phase rotators 107. Here, m=1, . . . ,Nc, mdn=1, . . . , N_(DM), and ndop_code(ndm)=1, . . . ,N_(DOP_CODE)(ndm).

Pseudo random noise (PN) codes, M-sequence codes, or Gold codes may beused as pseudo-random codes, for example. Pseudo-random code sequenceRCode is constituted by, for example, LRC code elements, as given byfollowing equation 53.

RCode={RC(1),RC(2), . . . ,RC(N _(LRC))}  (Equation 53)

Code elements of a pseudo-random code sequence include, for example,values of 1 to −1, where a phase rotation of 0 is applied for 1 and aphase rotation of π is applied for −1. A pseudo-random code sequence hascode length N_(LRC) less than or equal to N_(c). Further, for example,random code applier 111 cyclically varies code element index RC_INDEX(m)for the pseudo-random code sequence every m-th transmission period, asgiven by following equation 54.

[53]

RC_INDEX(m)=mod(m−1,N _(LRC))+1  (Equation 54)

Further, random code applier 111 outputs random code elementRC(RC_INDEX(m)) of pseudo-random code sequence RCode to random codemultiplier 214.

In radar transmitter 100 b, each of Nt phase rotators 107 applies, foreach transmission period Tr, coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm)(m) to which code element RC(RC_INDEX(m)) ofpseudo-random code sequence RCode is applied, namely, phase rotationamount ψ_(ndop_code(ndm), ndm)(m) angle[RC(RC_INDEX(m))], to a chirpsignal output from radar transmission signal generator 101.

For example, a description will be given of a case where when the numberof transmit antennas used for multiplexing transmission Nt=3, the numberof Doppler multiplexes N_(DM)=2, N_(CM)=2, orthogonal code sequencesCode₁={1, 1} and Code₂={1, −1} with code length Loc=2 are used, and thenumbers of coded Doppler multiplexes are set such that N_(DOP_CODE)(1)=1and N_(DOP_CODE)(2)=2.

In this case, random code applier 111 applies random codeRC(RC_INDEX(m)) to coded Doppler phase rotation amounts 111,ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) and outputs the resultingphase rotation amounts to phase rotators 107 for each transmissionperiod.

Phase rotator PROT #[1, 1] applies, for each transmission period, phaserotation to a chirp signal generated in radar transmission signalgenerator 101 in a manner given by following expression 55 for eachtransmission period. The output of phase rotator PROT #[1, 1] is outputfrom transmit antenna Tx #[1, 1]. Here, cp(t) denotes a chirp signal foreach transmission period.

[54]

exp[j{ψ _(1,1)(1)+angle[RC(1)]}]cp(t),exp[j{ψ_(1,1)(2)+angle[RC(2)]}]cp(t), . . . ,   (Expression 55)

Likewise, phase rotator PROT #[1, 2] applies, for each transmissionperiod, phase rotation to a chirp signal generated in radar transmissionsignal generator 101 in a manner given by following expression 56 foreach transmission period. The output of phase rotator PROT #[1, 2] isoutput from transmit antenna Tx #[1, 2].

[55]

exp[j{ψ _(1,2)(1)+angle[RC(1)]}]cp(t),exp[j{ψ_(1,2)(2)+angle[RC(2)]}]cp(t), . . . ,   (Expression 56)

Likewise, phase rotator PROT #[2, 2] applies, for each transmissionperiod, phase rotation to a chirp signal generated in radar transmissionsignal generator 101 in a manner given by following expression 57 foreach transmission period. The output of phase rotator PROT #[2, 2] isoutput from transmit antenna Tx #[2, 2].

[56]

exp[j{ψ _(2,2)(1)+angle[RC(1)]}]cp(t),exp[j{ψ_(2,2)(2)+angle[RC(2)]}]cp(t), . . . ,   (Expression 57)

In radar receiver 200 b, random code multiplier 214 multiplies outputsignal RFT_(z)(f_(b), m) of beat frequency analyzer 208 in transmissionperiod m by random code element RC(RC_INDEX(m)) input from random codeapplier 111. Random code multiplier 214 outputs a signal represented byRC(RC_INDEX(m))×RFT_(z)(f_(b), m) to output switching section 209. Here,z=1, . . . , Na.

With the operation described above, even in the presence of interferencefrom a plurality of radar apparatuses for which the same or somefrequency bands overlap, in radar apparatus 10 b, the interferencesignals can be converted into pseudo-random signals by random codemultiplier 214 before being input to Doppler analyzers 210. Accordingly,in the outputs of Doppler analyzers 210, the effect of diffusing thesignal power of interference waves over the Doppler frequency domain isachieved. For example, multiplication of a pseudo-random code sequencecan reduce peak power of an interference wave to about 1/Ncode. This cangreatly reduce the probability of CFAR section 211 in the subsequentstage erroneously detecting peaks of interference waves.

If phase rotators 107 do not have sufficient phase rotation accuracy, aphase rotation error caused by the application of a random code during atransmission period of an orthogonal code may cause interference betweenorthogonal codes. To address the interference, for example, code lengthN_(LRC) of the pseudo-random code sequence is set to be less than orequal to Ncode, and random code applier 111 may apply the same randomcode element during the transmission period of orthogonal code lengthLoc. For example, random code applier 111 may set pseudo-random codeelement index RC_INDEX(m) given by following equation 58 for eachtransmission period m.

$\begin{matrix}\lbrack 57\rbrack &  \\{{{RC\_ Index}(m)} = {{{mod}( {\frac{{floor}\lbrack {m - 1} \rbrack}{Loc},N_{LRC}} )} + 1}} & ( {{Equation}58} )\end{matrix}$

This allows the same random code element to be applied during thetransmission period of an orthogonal code. Even when phase rotators 107do not have sufficient phase rotation accuracy, phase rotation errorscaused by random codes are constant, and interference between orthogonalcodes can be reduced.

For example, a description will be given of a case where whenRC_INDEX(m) given by equation 58 is used, the number of transmitantennas used for multiplexing transmission Nt=3, the number of Dopplermultiplexes N_(DM)=2, N_(CM)=2, orthogonal code sequences Code₁={1, 1}and Code₂={1, −1} with code length Loc=2 are used, and the numbers ofcoded Doppler multiplexes are set such that N_(DOP_CODE)(1)=1 andN_(DOP_CODE)(2)=2.

In this case, random code applier 111 applies random codeRC(RC_INDEX(m)) to coded Doppler phase rotation amounts ψ_(1, 1)(m),ψ_(1, 2)(m), and ψ_(2, 2)(m) and outputs the resulting phase rotationamounts to phase rotators 107 for each transmission period.

Phase rotator PROT #[1, 1] applies, for each transmission period, phaserotation to a chirp signal generated in radar transmission signalgenerator 101 in a manner given by following expression 59 for eachtransmission period. The output of phase rotator PROT #[1, 1] is outputfrom transmit antenna Tx #[1, 1]. Here, cp(t) denotes a chirp signal foreach transmission period.

[58]

exp[j{ψ _(1,1)(1)+angle[RC(1)]}]cp(t),exp[j{ψ_(1,1)(2)+angle[RC(1)]}]cp(t),exp[j{ψ_(1,1)(3)+angle[RC(2)]}]cp(t),exp[j{ψ _(1,1)(4)+angle[RC(2)]}]cp(t), . .. ,   (Expression 59)

Likewise, phase rotator PROT #[1, 2] applies, for each transmissionperiod, phase rotation to a chirp signal generated in radar transmissionsignal generator 101 in a manner given by following expression 60 foreach transmission period. The output of phase rotator PROT #[1, 2] isoutput from transmit antenna Tx #[1, 2].

[59]

exp[j{ψ _(1,2)(1)+angle[RC(1)]}]cp(t),exp[j{ψ_(1,2)(2)+angle[RC(1)]}]cp(t),exp[j{ψ_(1,2)(3)+angle[RC(2)]}]cp(t),exp[j{ψ _(1,2)(4)+angle[RC(2)]}]cp(t), . .. ,   (Expression 60)

Likewise, phase rotator PROT #[2, 2] applies, for each transmissionperiod, phase rotation to a chirp signal generated in radar transmissionsignal generator 101 in a manner given by following expression 61 foreach transmission period. The output of phase rotator PROT #[2, 2] isoutput from transmit antenna Tx #[2, 2].

[60]

exp[j{ψ _(2,2)(1)+angle[RC(1)]}]cp(t),exp[j{ψ_(2,2)(2)+angle[RC(1)]}]cp(t),exp[j{ψ_(2,2)(3)+angle[RC(2)]}]cp(t),exp[j{ψ _(2,2)(4)+angle[RC(2)]}]cp(t), . .. ,   (Expression 61)

(Variation 6 of Embodiment 1)

Embodiment 1 has described the case where the output of phase rotatorPROT #[ndop_code(ndm), ndm] is transmitted from transmit antenna Tx#[ndop_code(ndm), ndm]; however, this is not required.

For example, the association between each of plural transmit antennas108 and the assignment of a coded Doppler multiplexed signal (in otherwords, a combination of Doppler shift amount and code sequence) may beset to vary for each frame in which a radar transmission signal istransmitted.

For example, when continuously performing radar position measurement,radar apparatus 10 may vary transmit antenna 108, which transmits theoutput of phase rotator PROT #[ndop_code(ndm), ndm], for each radarposition measurement (for example, every Nc transmission periods(Nc×Tr)). Here, ndm=1, . . . , N_(DM), and ndop_code(ndm)=1, . . . ,N_(DOP_CODE)(ndm).

For example, radar apparatus 10 may hold a plurality of assignmenttables for assignment as to from which of Nt transmit antennas #1, . . ., and #Nt to transmit the output of each of Nt phase rotators PROT#[ndop_code(mdm), ndm]. For example, radar apparatus 10 can modify anassignment table for each radar position measurement (for example, everyNc transmission periods (Nc×Tr)) to variably set transmit antenna 108 tobe used for transmission for each radar position measurement.

Accordingly, when continuously performing radar position measurement,radar apparatus 10 variably sets, for the output of each of Nt phaserotators 107 in each radar position measurement, the corresponding oneof transmit antennas 108. Accordingly, in cases such as when signalsaffected by interference (for example, inter-code interference)different depending on transmit antenna 108 are received, the effect ofrandomizing the influence of interference can be achieved by varyingtransmit antennas 108.

(Variation 7 of Embodiment 1)

In Embodiment 1, coder 106 applies phase rotation based on one or aplurality of orthogonal code sequences less than or equal to N_(CM) toeach of phase rotation amounts ϕ₁, . . . , and ϕ_(NDM) for applyingN_(DM) Doppler shift amounts output from Doppler shift setter 105 to setcoded Doppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m) andoutputs coded Doppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m)to phase rotators 107. However, the processing performed by coder 106 isnot limited to this.

Variation 7 describes a case where associations between transmitantennas 108 and Doppler multiplexed signals are set to vary for eachtransmission period.

For example, the assignment of Doppler multiplexing (in other words,Doppler shift amounts) to transmit antennas 108 may vary for eachtransmission period without changing the number of Doppler multiplexesused for coded Doppler multiplexing transmission.

For example, coder 106 sets, for each code element of an orthogonal codesequence (for example, each transmission period Tr), a coded Dopplerphase rotation amount using a phase rotation amount for applying adifferent Doppler shift amount. In other words, coder 106 may make thevalue of Doppler shift amount DOP_(ndm) different for each code elementof an orthogonal code sequence (for example, each transmission periodTr) in coded Doppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m)applied to a radar transmission signal to be transmitted from eachtransmit antenna 108.

The setting of coded Doppler phase rotation amounts in the mannerdescribed above achieves effects similar to those of Embodiment 1.Further, for example, when signals affected by interference (forexample, inter-code interference) different depending on transmitantenna 108 are received, the effect of randomizing the influence ofinterference can be achieved by varying transmit antennas 108.

For example, coder 106 may set coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm)(m) using following equation 62 instead ofequation 5.

$\begin{matrix}\lbrack 61\rbrack &  \\{{\psi_{{{ndop\_ code}{({ndm})}},{ndm}}(m)} = {{{{floor}\lbrack \frac{( {m - 1} }{Loc} \rbrack} \times \phi_{{{mod}({{{ndm} + {OC\_ INDEX} - 2},N_{DM}})} + 1}} + {{angle}\lbrack {{OC}_{{ndop\_ code}{({ndm})}}({OC\_ INDEX})} \rbrack}}} & ( {{Equation}62} )\end{matrix}$

In coded Doppler phase rotation amount ψ_(ndop_code(ndm), ndm)(m) givenby equation 62, a phase rotation amount for applying a Doppler shiftamount is set to vary for transmission periods in such manner as to beϕ_(mod(ndm+OC_INDEX-2,NDM)+1) for the duration of Loc transmissionperiods, the number of which is equal to the code length used for coding(the first term in equation 62), and the phase rotation amounts of Loccode elements OC_(ndop_code(ndm))(1), . . . , andOC_(ndop_code(ndm))(Loc) of code Code_(ndop_code(ndm)) used for codingare applied (the second term in equation 62). In equation 62, phaserotation amount ϕ_(mod(ndm+OC_INDEX-2,NDM)+1) for applying a Dopplershift amount is set to vary for each code element (for example,OC_INDEX).

As an example, a description will be given of a case where when Nt=3,N_(DM)=2, and N_(CM)=2, coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm1)(m) given by equation 62 is set.

In this case, the assignment of Doppler shift amounts DOP₁ and DOP₂ andorthogonal codes Code₁ and Code₂ is determined in accordance with, forexample, the setting of N_(DOP_CODE)(1) and N_(DOP_CODE)(2), asillustrated in FIGS. 17A and 17B. In FIGS. 17A and 17B, the horizontalaxis represents a combination of a Doppler shift amount used for thefirst code element (for example, OC_INDEX=1) and a Doppler shift amountused for the second code element (for example, OC_INDEX=2).

As illustrated in FIGS. 17A and 17B, for example, in a code sequencewith the code length for N_(CM)=2, the Doppler shift amountcorresponding to the first code element and the Doppler shift amountcorresponding to the second code element are different.

For example, a description will be given of a case where in coder 106,using equation 62, when the number of transmit antennas used formultiplexing transmission Nt=3, the number of Doppler multiplexesN_(DM)=2, N_(CM)=2, and orthogonal code sequences Code₁={1, 1} andCode₂={1, −1} with code length Loc=2 are used, the numbers of codedDoppler multiplexes are set such that N_(DOP_CODE)(1)=1 andN_(DOP_CODE)(2)=2 (see, for example, FIG. 17B). In this case, coder 106sets coded Doppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), andψ_(2, 2)(m) given by following equations 63 to 65 and outputs codedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m)to phase rotators 107.

[62]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }{0,0,ϕ₁,ϕ₂,2ϕ₁,2ϕ₂,3ϕ₁,3ϕ₂, . . . }  (Equation 63)

[63]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }{0,0,ϕ₂,ϕ₁,2ϕ₂,2ϕ₁,3ϕ₂,3ϕ₁, . . . }  (Equation 64)

[64]

{ψ_(2,2)(1),ψ_(2,2)(2),ψ_(2,2)(3),ψ_(2,2)(4),ψ_(2,2)(5),ψ_(2,2)(6),ψ_(2,2)(7),ψ_(2,2)(8),. . . }{0,π,ϕ₂,ϕ₁+π,2ϕ₂,2ϕ₁+π,3ϕ₂,3ϕ₁+π, . . . }  (Equation 65)

In each of coded Doppler phase rotation amounts ψ_(1, 1)(m),ψ_(1, 2)(m), and ψ_(2, 2)(m) given by equations 63 to 65, phase rotationamounts ϕ₁ and ϕ₂ for applying Doppler shift amounts DOP₁ and DOP₂ arealternately used in a period with code length Loc=2.

For ψ_(1, 2)(m) and ψ_(2, 2)(m) given by equations 64 and 65, phaserotations using orthogonal code sequences Code₁ and Code₂ are applied,with a relationship maintained in which phase rotation amounts ϕ₁ and ϕ₂for applying Doppler shift amounts DOP₁ and DOP₂ are the same phaserotation amount in a period with code length Loc=2. In other words, forψ_(1, 2)(m) and ψ_(2, 2)(m), phase rotation amounts for applying Dopplershift amounts are changed in a similar manner in a period with codelength Loc, and a plurality of orthogonal code sequences are used toperform code multiplexing.

For ψ_(1, 1)(m) and ψ_(1, 2)(m) given by equations 63 and 64, phaserotation amounts ϕ₁ and ϕ₂ for applying Doppler shift amounts DOP₁ andDOP₂ are different phase rotation amounts in a period with code lengthLoc=2. Also for ψ_(1, 1)(m) and ψ_(2, 2)(m) given by equations 63 and65, phase rotation amounts ϕ₁ and ϕ₂ for applying Doppler shift amountsDOP₁ and DOP₂ are different phase rotation amounts in a period with codelength Loc=2.

Here, for example, a description will be given of a case where a phaserotation amount for applying Doppler shift amount DOP_(ndm) isϕ_(ndm)=2π(ndm−1)/N_(DM) given by equation 1, and phase rotation amountϕ₁ for applying Doppler shift amount DOP₁, which is equal to 0, andphase rotation amount ϕ₂ for applying Doppler shift amount DOP₂, whichis equal to π, are used. In this case, coder 106 sets coded Dopplerphase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) givenby following equations 66 to 68 and outputs coded Doppler phase rotationamounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) to phase rotators 107.Here, m=1, . . . , Nc.

[65]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }{0,0,0,π,0,0,0,π, . . . }  (Equation 66)

[66]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }{0,0,π,0,0,0,π,0, . . . }  (Equation 67)

[67]

{ψ_(2,2)(1),ψ_(2,2)(2),ψ_(2,2)(3),ψ_(2,2)(4),ψ_(2,2)(5),ψ_(2,2)(6),ψ_(2,2)(7),ψ_(2,2)(8),. . . }{0,π,π,π,0,π,π,π, . . . }  (Equation 68)

As another example, a description will be given of a case where whenNt=6, N_(DM)=4, and N_(CM)=2, coded Doppler phase rotation amountψ_(ndop_code(ndm), ndm1)(m) given by equation 62 is set.

In this case, the assignment of Doppler shift amounts DOP₁, DOP₂, DOP₃,and DOP₄ and orthogonal codes Code₁ and Code₂ is determined inaccordance with, for example, the setting of N_(DOP_CODE)(1),N_(DOP_CODE)(2), N_(DOP_CODE)(3), and N_(DOP_CODE)(4), as illustrated inFIGS. 18A and 18B. In FIGS. 18A and 18B, the horizontal axis representsa combination of a Doppler shift amount used for the first code element(for example, OC_INDEX=1) and a Doppler shift amount used for the secondcode element (for example, OC_INDEX=2).

As illustrated in FIGS. 18A and 18B, for example, in a code sequence ofthe code length for N_(CM)=2, the Doppler shift amount corresponding tothe first code element and the Doppler shift amount corresponding to thesecond code element are different.

For example, a description will be given of a case where in coder 106,using equation 62, when the number of transmit antennas used formultiplexing transmission Nt=6, the number of Doppler multiplexesN_(DM)=4, N_(CM)=2, and orthogonal code sequences Code₁={1, 1} andCode₂={1, −1} with code length Loc=2 are used, the numbers of codedDoppler multiplexes are set such that N_(DOP_CODE)(1)=1,N_(DOP_CODE)(2)=1, N_(DOP_CODE)(3)=2, and N_(DOP_CODE)(4)=2. In thiscase, coder 106 sets coded Doppler phase rotation amounts ψ_(1, 1)(m),ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m)given by following equations 69 to 74 and outputs coded Doppler phaserotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m),ψ_(1, 4)(m), and ψ_(2, 4)(m) to phase rotators 107. Here, m=1, . . . ,Nc.

[68]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }{0,0,ϕ₁,ϕ₂,2ϕ₁,2ϕ₂,3ϕ₁,3ϕ₂, . . . }  (Equation 69)

[69]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }{0,0,ϕ₂,ϕ₃,2ϕ₂,2ϕ₃,3ϕ₂,3ϕ₃, . . . }  (Equation 70)

[70]

{ψ_(1,3)(1),ψ_(1,3)(2),ψ_(1,3)(3),ψ_(1,3)(4),ψ_(1,3)(5),ψ_(1,3)(6),ψ_(1,3)(7),ψ_(1,3)(8),. . . }{0,0,ϕ₃,ϕ₄,2ϕ₃,2ϕ₄,3ϕ₃,3ϕ₄, . . . }  (Equation 71)

[71]

{ψ_(2,3)(1),ψ_(2,3)(2),ψ_(2,3)(3),ψ_(2,3)(4),ψ_(2,3)(5),ψ_(2,3)(6),ψ_(2,3)(7),ψ_(2,3)(8),. . . }{0,π,ϕ₃,ϕ₄+π,2ϕ₃,2ϕ₄+π,3ϕ₃,3ϕ₄+π, . . . }  (Equation 72)

[72]

{ψ_(1,4)(1),ψ_(1,4)(2),ψ_(1,4)(3),ψ_(1,4)(4),ψ_(1,4)(5),ψ_(1,4)(6),ψ_(1,4)(7),ψ_(1,4)(8),. . . }{0,0,ϕ₄,ϕ₁,2ϕ₄,2ϕ₁,3ϕ₄,3ϕ₁, . . . }  (Equation 73)

[73]

{ψ_(2,4)(1),ψ_(2,4)(2),ψ_(2,4)(3),ψ_(2,4)(4),ψ_(2,4)(5),ψ_(2,4)(6),ψ_(2,4)(7),ψ_(2,4)(8),. . . }{0,π,ϕ₄,ϕ₁+π,2ϕ₄,2ϕ₁+π,3ϕ₄,3ϕ₁+π, . . . }  (Equation 74)

In each of coded Doppler phase rotation amountsψ_(ndop_code(ndm), ndm)(m) given by equations 69 to 74, phase rotationamounts ϕ_(ndm) and ϕ_(mod(ndm, NDM)+1) for applying Doppler shiftamounts DOP_(ndm) and DOP_(mod(ndm, NDM)+1) are used in a period withcode length Loc=2. In coded Doppler phase rotation amountsψ_(ndop_code(ndm), ndm)(m) with ndm being different, phase rotationamounts for applying Doppler shift amounts are different.

Further, for coded Doppler phase rotation amounts ψ_(1, ndm)(m), . . . ,and ψ_(n2dop_code(ndm), ndm)(m), phase rotations using orthogonal codesequences Code₁, . . . , and Code_(ndop_code(ndm)) are applied, with arelationship maintained in which phase rotation amounts ϕ_(ndm) andϕ_(mod(ndm, NDM)+1) for applying Doppler shift amounts DOP_(ndm) andDOP_(mod(ndm,NDM)+1) are the same phase rotation amount in a period withcode length Loc=2.

Here, for example, a description will be given of a case where a phaserotation amount for applying Doppler shift amount DOP_(ndm) isϕ_(ndm)=2π(ndm−1)/N_(DM) given by equation 1, and phase rotation amountϕ₁ for applying Doppler shift amount DOP₁, which is equal to 0, phaserotation amount ϕ₂ for applying Doppler shift amount DOP₂, which isequal to π/2, phase rotation amount ϕ₃ for applying Doppler shift amountDOP₃, which is equal to π, and phase rotation amount ϕ₄ for applyingDoppler shift amount DOP₄, which is equal to 3π/2, are used. In thiscase, coder 106 sets coded Doppler phase rotation amounts ψ_(1, 1)(m),ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m)given by following equations 75 to 80 and outputs coded Doppler phaserotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m),ψ_(1, 4)(m), and ψ_(2, 4)(m) to phase rotators 107. Here, m=1, . . . ,Nc.

$\begin{matrix}\lbrack 74\rbrack &  \\{\{ {{\psi_{1,1}(1)},{\psi_{1,1}(2)},{\psi_{1,1}(3)},{\psi_{1,1}(4)},{\psi_{1,1}(5)},{\psi_{1,1}(6)},{\psi_{1,1}(7)},{\psi_{1,1}(8)},\ldots} \} = \{ {0,0,0,\frac{\pi}{2},0,\pi,0,\frac{3\pi}{2},\ldots} \}} & ( {{Equation}75} )\end{matrix}$ $\begin{matrix}\lbrack 75\rbrack &  \\{\{ {{\psi_{1,2}(1)},{\psi_{1,2}(2)},{\psi_{1,2}(3)},{\psi_{1,2}(4)},{\psi_{1,2}(5)},{\psi_{1,2}(6)},{\psi_{1,2}(7)},{\psi_{1,2}(8)},\ldots} \} = \{ {0,0,\frac{\pi}{2},\pi,\pi,0,\frac{3\pi}{2},\pi,\ldots} \}} & ( {{Equation}76} )\end{matrix}$ $\begin{matrix}\lbrack 76\rbrack &  \\{\{ {{\psi_{1,3}(1)},{\psi_{1,3}(2)},{\psi_{1,3}(3)},{\psi_{1,3}(4)},{\psi_{1,3}(5)},{\psi_{1,3}(6)},{\psi_{1,3}(7)},{\psi_{1,3}(8)},\ldots} \} = \{ {0,0,\pi,\frac{3\pi}{2},0,\pi,\pi,\frac{\pi}{2},\ldots} \}} & ( {{Equation}77} )\end{matrix}$ $\begin{matrix}\lbrack 77\rbrack &  \\{\{ {{\psi_{2,3}(1)},{\psi_{2,3}(2)},{\psi_{2,3}(3)},{\psi_{2,3}(4)},{\psi_{2,3}(5)},{\psi_{2,3}(6)},{\psi_{2,3}(7)},{\psi_{2,3}(8)},\ldots} \} = \{ {0,\pi,\pi,\frac{\pi}{2},0,0,\pi,\frac{3\pi}{2},\ldots} \}} & ( {{Equation}78} )\end{matrix}$ $\begin{matrix}\lbrack 78\rbrack &  \\{\{ {{\psi_{1,4}(1)},{\psi_{1,4}(2)},{\psi_{1,4}(3)},{\psi_{1,4}(4)},{\psi_{1,4}(5)},{\psi_{1,4}(6)},{\psi_{1,4}(7)},{\psi_{1,4}(8)},\ldots} \} = \{ {0,0,\frac{3\pi}{2},0,\pi,0,\frac{\pi}{2},0,\ldots} \}} & ( {{Equation}79} )\end{matrix}$ $\begin{matrix}\lbrack 79\rbrack &  \\{\{ {{\psi_{2,4}(1)},{\psi_{2,4}(2)},{\psi_{2,4}(3)},{\psi_{2,4}(4)},{\psi_{2,4}(5)},{\psi_{2,4}(6)},{\psi_{2,4}(7)},{\psi_{2,4}(8)},\ldots} \} = \{ {0,\pi,\frac{3\pi}{2},\pi,\pi,\pi,\frac{\pi}{2},\pi,\ldots} \}} & ( {{Equation}80} )\end{matrix}$

Next, an example of the operation of radar receiver 200 when codedDoppler phase rotation amounts are set by coder 106 described above willbe described. In radar receiver 200, a code separation process performedby coded Doppler demultiplexer 212 is different from that in Embodiment1.

Coded Doppler demultiplexer 212 performs a code separation process onthe outputs of Doppler analyzers 210 in the z-th signal processor 206indicated by Doppler frequency indices (f_(s_comp_cfar)±(nfd−1)×ΔFD) ofN_(DM) coded Doppler multiplexed signals for distance indices f_(b_cfar)output from CFAR section 211.

The code separation process may be performed on, for example, all ofnfd=1, . . . , N_(DM) for all the candidates of DopCase=1, . . . ,N_(DM). Note that coded Doppler demultiplexer 212 detects a codedDoppler multiplexed signal for which the number of coded Dopplermultiplexes is set to be smaller than N_(CM), and performsdiscrimination of transmit antennas 108 and determination of a targetDoppler frequency. Accordingly, coded Doppler demultiplexer 212 performsa code separation process, as given by following equation 81, to reducethe operation amount of the separation process.

$\begin{matrix}\lbrack 80\rbrack &  \\{{{DeMUL}_{\mathcal{z}}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{noc} = 1}^{Loc}\lbrack {O{C_{ncm}^{*}( {noc} )}{{VFT}_{\mathcal{z}}^{noc}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{Dop}_{{{mod}({{{{ndm}1} + {noc} - 2},N_{DM}})} + 1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \text{ }\exp\{ {{- j}\frac{2{\pi( {f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times \Delta{FD}} - {DOP}_{1}} )}}{N_{code}} \times \frac{{noc} - 1}{Loc}} \}} \rbrack}} & ( {{Equation}81} )\end{matrix}$

The superscript asterisk (*) indicates the complex conjugate operator.Further, nfd=1, . . . , N_(DM), ncm=1, . . . , N_(CM), and DopCase=1, .. . , N_(DM).

For example, when a coded Doppler multiplexed signal for which thenumber of coded Doppler multiplexes is set to be smaller than N_(CM)uses coded Doppler phase rotation amount ψ_(ndop_code(ndm), ndm1)(m)given by equation 62, coded Doppler demultiplexer 212 performs the codeseparation process given by equation 81, with consideration given to theuse of DOP_(mod(ndm1+noc-2, NDM)+1) as the Doppler shift amount for thenoc-th code element.

In equation 81, coded Doppler demultiplexer 212 performs a codeseparation process using candidate DOPposi(DOP_(mod(ndm1+noc-2, NDM)+1),DopCase) including Doppler shift amount DOP_(mod(ndm1+noc-2, NDM)+1) forthe output of Doppler analyzer 210 for every noc-th code element withineach DopCase. Here, code separation signal DeMUL_(z) ^(ncm)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) given by equation81 represents a code separation signal that uses Code_(ncm) for distanceindex f_(b_cfar) and Doppler frequency index(f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) in the z-thsignal processor 206. The Doppler frequency index(f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) represents aDoppler frequency index in the output of Doppler analyzer 210 for thefirst code element when the coded Doppler phase rotation amount given byequation 62 is used.

In equation 81, in the exp term, since the sampling times for theoutputs of Doppler analyzers 210 for each code element are shifted,phase correction is performed in accordance with Doppler frequency index(f_(s_comp_cfar)+(DopCase−1)×ΔFD-DOP₁).

Further, Doppler shift amount DOP_(ndm) set in Doppler shift setter 105is represented by DOP₁<DOP₂< . . . <DOP_(DM-1)<DOP_(DM), and DOP₁ fallswithin the range of fs_comp=−Ncode/2, −Ncode/2+ΔFD−1 in the initialstate (when the relative velocity to the target is zero). Accordingly,coded Doppler demultiplexer 212 calculates an amount of phase correctionusing, for example, DOP₁ as a reference.

The subsequent processing performed by radar receiver 200 is similar tothat in Embodiment 1, and a description thereof is omitted here.

(Variation 8 of Embodiment 1)

Embodiment 1 has described a case where when radar apparatus 10repeatedly transmits chirp pulses Nc times as radar transmissionsignals, the center frequencies of the chirp signals are made constant(see, for example, FIG. 2 ). However, the center frequencies of thechirp signals are not necessarily constant.

Variation 8 describes a case where the center frequencies of chirpsignals are variably set.

[Configuration of Radar Apparatus]

FIG. 19 is a block diagram illustrating an example configuration ofradar apparatus 10 c according to Variation 8. In FIG. 19 , componentssimilar to those in Embodiment 1 (FIG. 1 ) are identified with the samenumerals, and a description thereof is omitted.

In the following, for example, radar apparatus 10 c transmits radartransmission signals such that center frequencies fc of chirp signalsare changed by Δf (for example, increased in a case where Δf>0 anddecreased in a case where Δf<0) for every transmission period Tr.

In radar transmitter 100 c, radar transmission signal generator 101 cincludes modulated signal emitter 102, VCO 103 c, and transmissionfrequency controller 112.

For example, modulated signal emitter 102 periodically emitssawtooth-shaped modulated signals for VCO control. Here, thetransmission period is represented by Tr.

Transmission frequency controller 112 controls, for each transmissionperiod Tr, center frequency fc of a frequency-modulated signal (chirpsignal) to be output from VCO 103 c. For example, transmission frequencycontroller 112 may change center frequency fc of a frequency-modulatedsignal by Δf for each transmission period Tr.

VCO 103 c outputs frequency-modulated signals to phase rotators 107 andradar receiver 200 c (for example, mixer section 204), based on theoutput of transmission frequency controller 112 and the output ofmodulated signal emitter 102.

FIG. 20 illustrates an example of frequency-modulated signals(hereinafter referred to as chirp signals).

In FIG. 20 , for example, VCO 103 c outputs a chirp signal having centerfrequency fc(1) equal to f₀ during the first transmission period Tr #1.Further, as illustrated in FIG. 20 , VCO 103 c outputs a chirp signalhaving center frequency fc(2) equal to f₀+Δf during the secondtransmission period Tr #2. Also in FIG. 20 , VCO 103 c outputs a chirpsignal having center frequency fc(m) equal to f₀+(m−1)Δf during the m-thtransmission period Tr #m. Accordingly, VCO 103 c changes the centerfrequency of a chirp signal by Δf for each transmission period Tr.

That is, in FIG. 20 , center frequency fc(N_(c)) of the chirp signal inthe N_(c)-th transmission period Tr #N_(c) is given by f₀+Δf×(N_(c)−1).

The respective chirp signals may be, for example, chirp signals havingthe same frequency-modulation bandwidth Bw in time width T_(A) of arange gate. In the example illustrated in FIG. 20 , the case where Δf>0(in other words, the case where center frequency fc is increased) isillustrated. The same applies to the case where Δf<0 (in other words,the case where center frequency fc is decreased).

Other operation of radar transmitter 100 c illustrated in FIG. 19 may besimilar to that in Embodiment 1.

Next, an example of the operation of radar receiver 200 c of radarapparatus 10 c will be described.

In radar receiver 200 c, the processing performed by antenna channelprocessors 201 on signals received by receive antennas 202, and theoperation of subsequent CFAR section 211 and coded Doppler demultiplexer212 are similar to the operation in Embodiment 1. In radar receiver 200c, furthermore, a direction estimation process performed by directionestimator 213 c using the output of coded Doppler demultiplexer 212 isalso similar to the operation in Embodiment 1.

In radar receiver 200 c, for example, a conversion process for Dopplervelocity information of the target based on the Doppler frequencydetermination result for the target (for example, Doppler frequencyf_(TARGET) of the target), which is performed by direction estimator 213c, is different from that in Embodiment 1.

The conversion of distance information R(f_(b)) based on the beatfrequency index is similar to that in Embodiment 1, and directionestimator 213 c may output, for example, based on equation 31, distanceinformation R(f_(b)) using beat frequency index (or distance index)f_(b).

Direction estimator 213 c may output Doppler velocity information v_(d)of the target detected in the following way using, for example, Dopplerfrequency f_(TARGET) of the target and distance index f_(b_cfar).

For example, when radar transmission signals for which centerfrequencies fc of chirp signals are changed by Δf for every transmissionperiod Tr are used, even if the relative velocity of the target is zero,center frequencies fc of chirp signals are changed for everytransmission period Tr. Accordingly, a reception signal of radarapparatus 10 c includes phase rotation caused by a change in the centerfrequency of a chirp signal for each transmission period Tr.

Center frequency fc in the m-th transmission period Tr for targetdistance R_(target) is changed by (m−1)Δf relative to the centerfrequency in the first transmission period Tr as a reference. Phaserotation amount Δη(m, R_(target)) caused by the change in centerfrequency fc is given by equation 81-1, with consideration given toreflected-wave arrival time (2R_(target)/C_(o)) from target distanceR_(target). Equation 81-1 indicates the relative phase rotation amountobtained using the phase of the first transmission period Tr as areference. C₀ denotes the speed of light

$\begin{matrix}\lbrack 81\rbrack &  \\{{{\Delta\eta}( {m,R_{target}} )} = {2{\pi( {m - 1} )}\Delta f \times ( \frac{2R_{target}}{C_{0}} )}} & ( {{Equation}81 - 1} )\end{matrix}$

Therefore, as given by following equation 81-2, direction estimator 213c calculates Doppler velocity information v_(d)(f_(TARGET), f_(b_cfar)),based on a transformation equation taking into account Δf that is theamount of change in center frequency fc of a chirp signal for eachtransmission period Tr.

$\begin{matrix}\lbrack 82\rbrack &  \\{{v_{d}( {f_{TARGET},f_{b\_ cfar}} )} = {\frac{C_{0}}{2f_{0}}( {f_{TARGET} - \frac{\Delta f \times 2{R( f_{b\_ car} )}}{T_{r} \times C_{0}}} )}} & ( {{Equation}81 - 2} )\end{matrix}$

In equation 81-2, the first term corresponds to equation 49 and is arelative Doppler velocity component represented by Doppler frequencyf_(TARGET). The second term in equation 81-2 is a Doppler velocitycomponent that is generated by changing center frequency fc of a chirpsignal by Δf for each transmission period Tr. For example, as given byequation 81-2, direction estimator 213 c can calculate the true relativeDoppler velocity v_(d)(f_(TARGET), f_(b_cfar)) of the target by removingthe Doppler component in the second term from the first term. Here,R(f_(b_cfar)) denotes distance information (distance estimation value)calculated from beat frequency index f_(b_cfar) in accordance withequation 31.

The Doppler range of the target is assumed to be up to ±1/(2×Tr). Ifv_(d)(f_(TARGET), f_(b_cfar)) satisfies v_(d)(f_(TARGET),f_(b_cfar))<−C₀/(4f₀Tr), direction estimator 213 c may output detectedDoppler velocity information v_(d)(f_(TARGET), f_(b_cfar)) of the targetin accordance with following equation 81-3.

$\begin{matrix}\lbrack 83\rbrack &  \\{{v_{d}( {f_{TARGET},f_{b\_ cfar}} )} = {\frac{C_{0}}{2f_{0}}( {f_{TARGET} + \frac{1}{T_{r}} - \frac{\Delta f \times 2{R( f_{b\_ car} )}}{T_{r} \times C_{0}}} )}} & ( {{Equation}81 - 3} )\end{matrix}$

Also, since the Doppler range of the target is assumed to be up to±1/(2×Tr), if v_(d)(f_(TARGET), f_(b_cfar)) satisfies v_(d)(f_(TARGET),f_(b_cfar))>C₀/(4f₀ Tr), direction estimator 213 c may output detectedDoppler velocity information v_(d)(f_(TARGET), f_(b_cfar)) of the targetin accordance with following equation 81-4.

$\begin{matrix}{\lbrack 84\rbrack} &  \\{{v_{d}( {f_{TARGET},f_{b\_ cfar}} )} = {\frac{C_{0}}{2f_{0}}( {f_{TARGET} - \frac{1}{T_{r}} - \frac{\Delta f \times 2{R( f_{b\_ cfar} )}}{T_{r} \times C_{0}}} )}} & ( {{Equation}81 - 4} )\end{matrix}$

As described above, in Variation 8, in radar apparatus 10 c, centerfrequencies fc of chirp signals are changed based on transmissionperiods Tr of radar transmission signals. For example, radar apparatus10 c transmits radar transmission signals such that center frequenciesfc of chirp signals are changed by Δf (for example, increased in a casewhere Δf>0 and decreased in a case where Δf<0) for every transmissionperiod Tr. Even in this case, as in Embodiment 1, radar apparatus 10 c(for example, MIMO radar) can extend the effective Doppler frequencybandwidth to 1/(Tr) and can extend the Doppler frequency (relativevelocity) detection range with no ambiguity. In addition, radarapparatus 10 c can reduce mutual interference between multiplexedsignals to about the noise level. According to Variation 8, therefore,radar apparatus 10 c can improve target-object sensing accuracy over awider Doppler frequency range.

Variation 8 is applicable not only to Embodiment 1 but also toVariations 1 to 7 of Embodiment 1, and respective similar effects can beachieved. For example, in any of Variations 1 to 7 of Embodiment 1,radar transmission signals including chirp signals whose centerfrequencies fc are changed in the manner illustrated in FIG. 20 may beused.

In Variation 8, furthermore, for example, radar apparatus 10 c transmitsradar transmission signals in such a manner that center frequencies fcof chirp signals are changed by Δf for every transmission period Tr, andcan thus improve the distance resolution by a change width of the centerfrequencies of chirp signals (see, for example, NPL 4). According toVariation 8, since the distance resolution can be improved by a changewidth of the center frequencies of chirp signals, the chirp sweepbandwidth (for example, Bw) can be reduced compared with transmissionwith the center frequencies of chirp signals kept constant. Thereduction in chirp sweep bandwidth can reduce, for example, thetransmission period Tr while improving the distance resolution. As aresult, in code multiplexing transmission, the detectable Doppler rangewithout ambiguity can further be extended.

(Variation 9 of Embodiment 1)

The periods in which the center frequencies of chirp signals are changedare not limited to transmission periods Tr as in Variation 8. Variation9 describes a case where the center frequencies of chirp signals are setto vary in increments of transmission periods of a plurality of chirpsignals.

For example, Variation 9 describes a case where the center frequenciesof chirp signals are set to vary for every Loc transmission periods(Loc×Tr) (in other words, a transmission period of an orthogonal codesequence, hereinafter referred to as “code transmission period”), thenumber of which is equal to the code length of a single orthogonal codeused for Doppler multiplexing transmission.

[Configuration of Radar Apparatus]

FIG. 21 is a block diagram illustrating an example configuration ofradar apparatus 10 d according to Variation 9. In FIG. 21 , componentssimilar to those in Embodiment 1 (FIG. 1 ) or Variation 8 (FIG. 19 ) areidentified with the same numerals, and a description thereof is omitted.

In Variation 9, for example, radar apparatus 10 d transmits radartransmission signals such that center frequencies fc of chirp signalsare changed by Δf (for example, increased in a case where Δf>0 anddecreased in a case where Δf<0) for every code transmission period(Loc×Tr).

In radar transmitter 100 d, radar transmission signal generator 101 dincludes modulated signal emitter 102, VCO 103 d, and transmissionfrequency controller 112 d. In radar transmitter 100 d, phase rotationamount setter 104 d includes Doppler shift setter 105 and coder 106 d.

For example, in radar transmission signal generator 101 d, modulatedsignal emitter 102 periodically emits sawtooth-shaped modulated signalsfor VCO control. Here, the transmission period is represented by Tr.

Transmission frequency controller 112 d controls, based on orthogonalcode element index OC_INDEX output from coder 106 d, center frequency fcof a frequency-modulated signal (chirp signal) to be output from VCO 103d for each code transmission period (Loc×Tr).

For example, transmission frequency controller 112 d may change centerfrequency fc of a frequency-modulated signal to be output from VCO 103 dby Δf in transmission period Tr corresponding to OC_INDEX=1. In otherwords, transmission frequency controller 112 d controls, in transmissionperiod Tr corresponding to OC_INDEX≠1, center frequency fc of afrequency-modulated signal to be output from VCO 103 d to be the same ascenter frequency fc in previous transmission period Tr. With thiscontrol, transmission frequency controller 112 d can perform controlsuch that center frequencies fc are changed by Δf for every codetransmission period (Loc×Tr).

VCO 103 d outputs frequency-modulated signals to phase rotators 107 andradar receiver 200 d (for example, mixer section 204), based on theoutput of transmission frequency controller 112 d and the output ofmodulated signal emitter 102.

FIG. 22 illustrates an example of frequency-modulated signals(hereinafter referred to as chirp signals).

In FIG. 22 , for example, VCO 103 d outputs a chirp signal having centerfrequency fc(1) equal to f₀ during the first transmission period Tr #1(for example, OC_INDEX=1). Further, as illustrated in FIG. 22 , VCO 103d outputs a chirp signal having center frequency fc(2) equal to f₀during the second transmission period Tr #2 (for example, OC_INDEX=2).Likewise, VCO 103 d outputs chirp signals having center frequenciesfc(3) to fc(Loc) equal to f₀ during the third transmission period (forexample, OC_INDEX=3) (not illustrated) to the Loc-th transmission periodTr #Loc (for example, OC_INDEX=Loc), respectively.

VCO 103 d outputs a chirp signal having center frequency fc(Loc+1) equalto f₀+Δf during the (Loc+1)-th transmission period Tr #(Loc+1). Further,VCO 103 d outputs chirp signals having center frequencies fc(Loc+2) tofc(2Loc) equal to f₀+Δf during the (Loc+2)-th transmission period Tr#(Loc+2) to the (2Loc)-th transmission period Tr #(2Loc), respectively.

Likewise, VCO 103 d outputs a chirp signal having center frequency fc(m)equal to f₀+floor[(m−1)/Loc]Δf during the m-th transmission period Tr#m.

That is, in FIG. 22 , center frequency fc(N_(c)) of the chirp signal inthe N_(c)-th transmission period Tr #N_(c) is given by f₀+(Ncode−1)Δf.Here, Ncode=N_(c)/Loc.

The respective chirp signals may be, for example, chirp signals havingthe same frequency-modulation bandwidth Bw in time width T_(A) of arange gate. In the example illustrated in FIG. 22 , the case where Δf>0(in other words, the case where center frequency fc is increased) isillustrated. The same applies to the case where Δf<0 (in other words,the case where center frequency fc is decreased).

Other operation of radar transmitter 100 d illustrated in FIG. 21 may besimilar to that in Embodiment 1.

Next, an example of the operation of radar receiver 200 d of radarapparatus 10 d will be described.

In radar receiver 200 d, the processing performed by antenna channelprocessors 201 on signals received by receive antennas 202, and theoperation of subsequent CFAR section 211 are similar to the operation inEmbodiment 1. In radar receiver 200 d, furthermore, a directionestimation process performed by direction estimator 213 d using theoutput of coded Doppler demultiplexer 212 d is also similar to theoperation in Embodiment 1.

In radar receiver 200 d, for example, the operation of coded Dopplerdemultiplexer 212 d and a conversion process for Doppler velocityinformation of the target, which is performed by direction estimator 213d, is different from those in Embodiment 1.

The conversion of distance information R(f_(b)) based on the beatfrequency index is similar to that in Embodiment 1, and directionestimator 213 d may output, for example, based on equation 31, distanceinformation R(f_(b)) using beat frequency index (or distance index)f_(b).

The following describes an example of the operation of coded Dopplerdemultiplexer 212 d, which is different from that in Embodiment 1.

For example, when radar transmission signals for which centerfrequencies fc of chirp signals are changed by Δf for every codetransmission period (Loc×Tr) are used, even if the relative velocity ofthe target is zero, center frequencies fc of chirp signals are changedfor every code transmission period (Loc×Tr). Accordingly, the output ofeach of Loc Doppler analyzers 210 of radar apparatus 10 d includes phaserotation caused by a change in the center frequency of a chirp signalfor each code transmission period (Loc×Tr).

For example, center frequency fc in the m-th transmission period Tr fortarget distance R_(target) is changed by floor[(m−1)/Loc]Δf relative tocenter frequency fc in the first transmission period Tr as a reference.Accordingly, phase rotation amount Δη(m, R_(target)) caused by thechange in center frequency is given by equation 81-5, with considerationgiven to reflected-wave arrival time (2R_(target)/C₀) from targetdistance R_(target) Equation 81-5 indicates the relative phase rotationamount obtained using the phase of the first transmission period Tr as areference. C₀ denotes the speed of light.

$\begin{matrix}\lbrack 85\rbrack &  \\{{\Delta{\eta( {m,R_{target}} )}} = {2\pi{floor}( \frac{m - 1}{Loc} )\Delta f \times ( \frac{2R_{target}}{C_{0}} )}} & ( {{Equation}81 - 5} )\end{matrix}$

Each of Loc Doppler analyzers 210 performs Doppler analysis that takesinto account the phase rotation given by equation 81-5. Here, codetransmission period (Loc×Tr) in which center frequency fc of a chirpsignal is changed by Δf matches the period for switching Doppleranalyzers 210 for every Loc code elements. Accordingly, to correctDoppler phase rotation caused by the time difference of Doppler analysisamong Loc Doppler analyzers 210 to separate code-multiplexed signals,coded Doppler demultiplexer 212 d uses following equation 81-6 insteadof equation 35 described in <(1) code separation process> in Embodiment1.

$\begin{matrix}{\lbrack 86\rbrack} &  \\{{{DeMUL}_{z}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{noc} = 1}^{Loc}\lbrack {{{OC}_{ncm}^{*}({noc})}{{VFT}_{z}^{noc}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {{- j}\frac{2{\pi\begin{pmatrix}{f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times}} \\{{\Delta{FD}} - {{DO}P_{1}}}\end{pmatrix}}}{N_{code}} \times \frac{{noc} - 1}{Loc}} \}\exp\{ {j2{\pi\Delta}f\frac{2{R( f_{b\_ cfar} )}}{C_{0}}\frac{{noc} - 1}{Loc}} \}} \rbrack}} & ( {{Equation}81 - 6} )\end{matrix}$

Equation 81-6 is different from equation 35 in the additional term forcorrecting the phase rotation caused by the change in center frequency:

$\begin{matrix}\lbrack 87\rbrack &  \\{\exp\lbrack {j2{\pi\Delta}f\frac{2{R( f_{b\_ cfar} )}}{C_{0}}\frac{{noc} - 1}{Loc}} } & \end{matrix}$

Here, R(f_(b_cfar)) denotes a distance estimation value calculated frombeat frequency index f_(b_cfar) in accordance with equation 31.

In equation 81-6, due to the change of Δf in reflected-wave arrival time(2R(f_(b_cfar))/C_(o)) from R(f_(b_cfar)), the phase rotation amount isgiven by 2πΔf×(2R(f_(b_cfar))/C_(o)) in code transmission period(Loc×Tr). Therefore, phase rotation caused by the time difference ofDoppler analysis among Loc Doppler analyzers 210 is, for example, forthe noc-th Doppler analyzer 210, (noc−1)/Loc times relative to that forthe first Doppler analyzer 210. Equation 81-6 is derived from thissituation. Note that noc=1, . . . , Loc.

In this manner, the period in which center frequency fc of a chirpsignal is changed by Δf is made equal to code transmission period(Loc×Tr) (or transmission periods, the number of which is equal to adivisor of Loc). Accordingly, the period in which the center frequencyof a chirp signal is changed matches the period for switching Doppleranalyzers 210 for each code element. For example, the period in whichcenter frequency fc of a chirp signal is changed by Δf matches theperiod for switching Doppler analyzers 210 for each code element,thereby allowing coded Doppler demultiplexer 212 d to uniquely decide aphase correction value for each of Loc Doppler analyzers 210. Thisenables coded Doppler demultiplexer 212 d to easily perform phasecorrection in (1) code separation process. Therefore, coded Dopplerdemultiplexer 212 d can easily perform the code separation process.

Next, an example of the operation of direction estimator 213 d that isdifferent from that in Embodiment 1 (for example, a conversion processfor Doppler velocity information of the target) will be described.

Direction estimator 213 d may output Doppler velocity information v_(d)of the target detected in the following way using, for example, Dopplerfrequency f_(TARGET) of the target and distance index f_(b_cfar).

When radar transmission signals for which center frequencies fc of chirpsignals are changed by Δf for every code transmission period (Loc×Tr)are used, even if the relative velocity of the target is zero, centerfrequencies fc of chirp signals are changed for every code transmissionperiod (Loc×Tr). Accordingly, a reception signal of radar apparatus 10 dincludes phase rotation caused by a change in the center frequency of achirp signal for each code transmission period (Loc×Tr).

Center frequency fc in the m-th transmission period Tr for targetdistance R_(target) is changed by floor[(m−1)/Loc]Δf relative to thecenter frequency in the first transmission period Tr as a reference.Phase rotation amount Δη(m, R_(target)) caused by the change in centerfrequency fc is given by equation 81-7, with consideration given toreflected-wave arrival time (2R_(target)/C_(o)) from target distanceR_(target). Equation 81-7 indicates the relative phase rotation amountobtained using the phase of the first transmission period Tr as areference. C₀ denotes the speed of light

$\begin{matrix}\lbrack 88\rbrack &  \\{{\Delta{\eta( {m,R_{target}} )}} = {2\pi{floor}( \frac{m - 1}{Loc} )\Delta f \times ( \frac{2R_{target}}{C_{0}} )}} & ( {{Equation}81 - 7} )\end{matrix}$

Therefore, as given by following equation 81-8, direction estimator 213d calculates Doppler velocity information v_(d)(f_(TARGET), f_(b_cfar)),based on a transformation equation taking into account Δf that is theamount of change in center frequency fc of a chirp signal for each codetransmission period (Loc×Tr).

$\begin{matrix}{\lbrack 89\rbrack} &  \\{{v_{d}( {f_{TARGET},f_{b\_ cfar}} )} = {\frac{C_{0}}{2f_{0}}( {f_{TARGET} - \frac{\Delta f \times 2{R( f_{b\_ cfar} )}}{{Loc} \times T_{r} \times C_{0}}} )}} & ( {{Equation}81 - 8} )\end{matrix}$

In equation 81-8, the first term corresponds to equation 49 and is arelative Doppler velocity component represented by Doppler frequencyf_(TARGET). The second term in equation 81-8 is a Doppler velocitycomponent that is generated by changing center frequency fc of a chirpsignal by Δf for each code transmission period (Loc×Tr). For example, asgiven by equation 81-8, direction estimator 213 d can calculate the truerelative Doppler velocity v_(d)(f_(TARGET), f_(b_cfar)) of the target byremoving the Doppler component in the second term from the first term.Here, R(f_(b_cfar)) denotes distance information (distance estimationvalue) calculated from beat frequency index f_(b_cfar) in accordancewith equation 31.

The Doppler range of the target is assumed to be up to ±1/(2×Tr). Ifv_(d)(f_(TARGET), f_(b_cfar)) satisfies v_(d)(f_(TARGET),f_(b_cfar))<−C₀/(4f₀Tr), direction estimator 213 d may output detectedDoppler velocity information v_(d)(f_(TARGET), f_(b_cfar)) of the targetin accordance with following equation 81-9.

$\begin{matrix}{\lbrack 90\rbrack} &  \\{{v_{d}( {f_{es\_ cfar},f_{b\_ cfar}} )} = {\frac{C_{0}}{2f_{0}}( {f_{TARGET} + \frac{1}{T_{r}} - \frac{\Delta f \times 2{R( f_{b\_ cfar} )}}{{Loc} \times T_{r} \times C_{0}}} )}} & ( {{Equation}81 - 9} )\end{matrix}$

Also, since the Doppler range of the target is assumed to be up to±1/(2×Tr), if v_(d)(f_(TARGET), f_(b_cfar)) satisfies v_(d)(f_(TARGET),f_(b_cfar))>C₀/(4f₀ Tr), direction estimator 213 d may output detectedDoppler velocity information v_(d)(f_(TARGET), f_(b_cfar)) of the targetin accordance with following equation 81-10.

$\begin{matrix}{\lbrack 91\rbrack} &  \\{{v_{d}( {f_{es\_ cfar},f_{b\_ cfar}} )} = {\frac{C_{0}}{2f_{0}}( {f_{TARGET} - \frac{1}{T_{r}} - \frac{\Delta f \times 2{R( f_{b\_ cfar} )}}{{Loc} \times T_{r} \times C_{0}}} )}} & ( {{Equation}81 - 10} )\end{matrix}$

As described above, in Variation 9, in radar apparatus 10 d, centerfrequencies fc of chirp signals are changed based on code transmissionperiods (Loc×Tr). For example, radar apparatus 10 d transmits radartransmission signals such that center frequencies fc of chirp signalsare changed by Δf (for example, increased in a case where Δf>0 anddecreased in a case where Δf<0) for every code transmission period(Loc×Tr). Even in this case, as in Embodiment 1, radar apparatus 10 d(for example, MIMO radar) can extend the effective Doppler frequencybandwidth to 1/(Tr) and can extend the Doppler frequency (relativevelocity) detection range with no ambiguity. In addition, radarapparatus 10 d can reduce mutual interference between code-multiplexedsignals to about the noise level. According to Variation 9, therefore,radar apparatus 10 d can improve target-object sensing accuracy over awider Doppler frequency range.

Variation 9 is applicable not only to Embodiment 1 but also toVariations 1 to 7 of Embodiment 1, and respective similar effects can beachieved. For example, in any of Variations 1 to 7 of Embodiment 1,radar transmission signals including chirp signals whose centerfrequencies fc are changed in the manner illustrated in FIG. 22 may beused.

According to Variation 9, furthermore, when the period in which centerfrequency fc of a chirp signal is changed by Δf is a plurality oftransmission periods Tr, for example, the period in which centerfrequency fc of a chirp signal is changed is made to match codetransmission period (Loc×Tr). Accordingly, the period in which centerfrequency fc of a chirp signal is changed is also made to match theperiod for switching Doppler analyzers 210 for each code element (inother words, the transmission period of a code used for codemultiplexing). Therefore, coded Doppler demultiplexer 212 d can easilyperform phase correction in a code demultiplexing process.

In Variation 9, furthermore, radar apparatus 10 d transmits, forexample, radar transmission signals in such a manner that centerfrequencies fc of chirp signals are changed by Δf for every codetransmission period (Loc×Tr), with the change width of the centerfrequencies of chirp signals being given by Δf×Ncode and the distanceresolution being given by 0.5C₀/(Δf×Ncode). For example, as the valuegiven by Δf×Ncode increases, the distance resolution can be moreimproved by the change width of the center frequencies of chirp signals.According to Variation 9, since the distance resolution can be improvedby a change width of the center frequencies of chirp signals, the chirpsweep bandwidth (for example, Bw) can be reduced compared withtransmission with the center frequencies of chirp signals kept constant.The reduction in chirp sweep bandwidth can reduce, for example, thetransmission period Tr while improving the distance resolution. As aresult, in code multiplexing transmission, the detectable Doppler rangewithout ambiguity can further be extended.

While Variation 9 has described the case where radar transmissionsignals are used for which center frequencies fc of chirp signals arechanged by Δf for every code transmission period (Loc×Tr), the period inwhich center frequency fc of a chirp signal is changed is not limited tothis. For example, radar apparatus 10 d may use radar transmissionsignals for which center frequencies fc of chirp signals are changed byΔf for every transmission periods, the number of which is equal to adivisor of code length Loc (divisor of (Loc×Tr)). When 1 is used amongdivisors of code length Loc, as in Variation 8, center frequencies fcare changed by Δf for every transmission period Tr.

When radar transmission signals for which center frequencies fc of chirpsignals are changed by Δf for every transmission periods, the number ofwhich is equal to divisor E of code length Loc, that is, for every Etransmission periods (E×Tr), are used, center frequencies fc of chirpsignals are changed by an amount corresponding to Δf×Loc/ε for everycode transmission period (Loc×Tr). Thus, in this case, Δf in equations81-8, 81-9, and 81-10 may be replaced by Δf×Loc/ε. Further, codedDoppler demultiplexer 212 d uses following equation 81-11 instead ofequation 81-6 in the code separation process. Here, E denotes a divisorof Loc.

$\begin{matrix}{\lbrack 92\rbrack} &  \\{{{DeMUL}_{z}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{noc} = 1}^{Loc}\lbrack {O{C_{ncm}^{*}({noc})}{{VFT}_{z}^{noc}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {{- j}\frac{2{\pi\begin{pmatrix}{f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times}} \\{{\Delta{FD}} - {{DO}P_{1}}}\end{pmatrix}}}{N_{code}} \times \frac{{noc} - 1}{Loc}} \} \times \exp\{ {j2{\pi\Delta}f\frac{2{R( f_{b\_ cfar} )}}{C_{0}}\frac{{noc} - 1 - {{floor}( {( {{noc} - 1} )/\varepsilon} )}}{\varepsilon}} \}} \rbrack}} & ( {{Equation}81 - 11} )\end{matrix}$

(Variation 10 of Embodiment 1)

The method for controlling center frequency fc of a frequency-modulatedsignal (chirp signal) is not limited to the methods according toVariation 8 (FIG. 20 ) and Variation 9 (FIG. 22 ). Variation 10describes another method for controlling center frequency fc of afrequency-modulated signal (chirp signal).

The configuration of a radar apparatus according to Variation 10 may besimilar to, for example, the configuration of radar apparatus 10 daccording to Variation 9.

In Variation 10, for example, radar apparatus 10 d periodically changescenter frequency fc of a chirp signal over a plurality of transmissionperiods within code transmission period (Loc×Tr). In this case, thetiming at which center frequencies fc of chirp signals are changed in around is made to match code transmission period (Loc×Tr), therebyallowing radar apparatus 10 d to reduce the amount of signal processingperformed by coded Doppler demultiplexer 212 d (the details will bedescribed below).

For example, in radar transmitter 100 d, modulated signal emitter 102periodically emits sawtooth-shaped modulated signals for VCO control.Here, the transmission period is represented by Tr.

Transmission frequency controller 112 d controls, based on orthogonalcode element index OC_INDEX output from coder 106 d in phase rotationamount setter 104 d, center frequency fc of a frequency-modulated signal(chirp signal) output from VCO 103 d for each transmission period Tr.

For example, transmission frequency controller 112 d sets centerfrequency fc of a frequency-modulated signal to be output from VCO 103 dto f₀ in transmission period Tr corresponding to OC_INDEX=1. Further,transmission frequency controller 112 d sets center frequency fc of afrequency-modulated signal to be output from VCO 103 d to f₀+Δf intransmission period Tr corresponding to OC_INDEX=2. Likewise,transmission frequency controller 112 d sets center frequencies fc ofthe frequency-modulated signals to be output from VCO 103 d to (f₀+2Δf)to (f₀+(Loc−1)Δf) in transmission periods Tr where OC_INDEX=3 to Loc,respectively.

With this control, transmission frequency controller 112 d can performcontrol such that, for example, center frequencies fc of chirp signalsare periodically changed over every code transmission period (Loc×Tr).

VCO 103 d outputs frequency-modulated signals to phase rotators 107 andradar receiver 200 d (for example, mixer section 204), based on theoutput of transmission frequency controller 112 d and the output ofmodulated signal emitter 102.

FIG. 23 illustrates an example of frequency-modulated signals(hereinafter referred to as chirp signals).

In FIG. 23 , for example, VCO 103 d outputs a chirp signal having centerfrequency fc(1) equal to f₀ during the first transmission period Tr #1.Further, for example, VCO 103 d outputs a chirp signal having centerfrequency fc(2) equal to f₀+Δf during the second transmission period Tr#2. Likewise, VCO 103 d outputs chirp signals having center frequenciesfc(3) to fc(Loc) equal to (f₀+240 to (f₀+(Loc−1)Δf) during the thirdtransmission period Tr #3 to the Loc-th transmission period Tr #Loc,respectively.

Further, VCO 103 d outputs a chirp signal having center frequencyfc(Loc+1) equal to f₀ during the (Loc+1)-th transmission period Tr#(Loc+1). Likewise, VCO 103 d outputs chirp signals having centerfrequencies fc(Loc+2) to fc(2Loc) equal to (f₀+Δf) to (f₀+(Loc−1)Δf)during the (Loc+2)-th transmission period Tr #(Loc+2) to the (2Loc)-thtransmission period Tr #(2Loc), respectively.

Likewise, VCO 103 d outputs a chirp signal having center frequency fc(m)equal to f₀+mod(m−1, Loc)Δf during the m-th transmission period Tr #m.

That is, in FIG. 23 , center frequency fc(N_(c)) of the chirp signal inthe N_(c)-th transmission period Tr #N_(c) is given by f₀+(Loc−1)Δf.

In this manner, in FIG. 23 , center frequencies fc of chirp signals arechanged in a round in a period that is a divisor multiple of the codelength of a code sequence (for example, orthogonal code) relative to thetransmission period of the radar transmission signal. In other words,for example, in FIG. 23 , center frequencies fc of chirp signals are thesame in transmission periods Tr having the same OC_INDEX.

The respective chirp signals may be, for example, chirp signals havingthe same frequency-modulation bandwidth Bw in time width TA of a rangegate. In the example illustrated in FIG. 23 , the case where Δf>0 (inother words, the case where center frequency fc is increased) isillustrated. The same applies to the case where Δf<0 (in other words,the case where center frequency fc is decreased).

Other operation of radar transmitter 100 d illustrated in FIG. 21 may besimilar to that in Embodiment 1.

Next, an example of the operation of radar receiver 200 d of radarapparatus 10 d will be described.

In radar receiver 200 d, the processing performed by antenna channelprocessors 201 on signals received by receive antennas 202, and theoperation of subsequent CFAR section 211 are similar to the operation inEmbodiment 1. In radar receiver 200 d, furthermore, a directionestimation process performed by direction estimator 213 d using theoutput of coded Doppler demultiplexer 212 d is also similar to theoperation in Embodiment 1.

In radar receiver 200 d, for example, the operation of coded Dopplerdemultiplexer 212 d and a conversion process for Doppler velocityinformation of the target, which is performed by direction estimator 213d, is different from that in Embodiment 1.

The conversion of distance information R(f_(b)) based on the beatfrequency index is similar to that in Embodiment 1, and directionestimator 213 d may output, for example, based on equation 31, distanceinformation R(f_(b)) using beat frequency index (or distance index)f_(b).

The following describes an example of the operation of coded Dopplerdemultiplexer 212 d, which is different from that in Embodiment 1.

For example, when radar transmission signals for which centerfrequencies fc of chirp signals are changed by f₀, f₀+Δf, f₀+(Loc−1)Δffor every transmission period Tr within each code transmission period(Loc×Tr) are used, radar reflected waves obtained in response to thetransmission of chirp signals whose center frequencies fc are f₀, f₀+Δf,and f₀+(Loc−1)Δf are input to the first, second, . . . , and Loc-thDoppler analyzers 210 as reception signals.

Thus, the center frequencies of the respective radar reflected waves tobe input to Loc Doppler analyzers 210 are the same. Accordingly, LocDoppler analyzers 210 perform Doppler analysis on reception signals thatare radar reflected waves obtained when chirp signals having the samecenter frequency fc are transmitted.

In contract, center frequencies fc of the chirp signals differ in LocDoppler analyzers 210. Accordingly, to correct phase rotation caused bythe time difference of Doppler analysis among Loc Doppler analyzers 210to separate code-multiplexed signals, coded Doppler demultiplexer 212 duses following equation 81-12 instead of equation 35 described in <(1)code separation process> in Embodiment 1.

$\begin{matrix}{\lbrack 93\rbrack} &  \\{{{DeMUL}_{z}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{noc} = 1}^{Loc}\lbrack {O{C_{ncm}^{*}({noc})}{{VFT}_{z}^{noc}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {{- j}\frac{2{\pi\begin{pmatrix}{f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times}} \\{{\Delta{FD}} - {{DO}P_{1}}}\end{pmatrix}}}{N_{code}} \times \frac{{noc} - 1}{Loc}} \}\exp\{ {{- j}2{\pi( {{noc} - 1} )}\Delta f\frac{2{R( f_{b\_ cfar} )}}{C_{0}}} \}} \rbrack}} & ( {{Equation}81 - 12} )\end{matrix}$

Equation 81-12 is different from equation 35 in the additional term forcorrecting the phase rotation caused by the change in center frequency:

$\begin{matrix}\lbrack 94\rbrack &  \\{\exp\lbrack {{- j}2{\pi( {{noc} - 1} )}\Delta f\frac{2{R( f_{b\_ cfar} )}}{C_{0}}} \rbrack} & \end{matrix}$

Here, R(f_(b_cfar)) denotes a distance estimation value calculated frombeat frequency index f_(b_cfar) in accordance with equation 31.

Equation 81-12 is derived from the following. For example, when thecenter frequency of a transmission chirp signal for the first Doppleranalyzer 210 is used as a reference, the center frequencies oftransmission chirp signals for the second to Loc-th Doppler analyzers210 are different by Δf, . . . , and (Loc−1)Δf, respectively.Accordingly, phase rotation amounts in reflected-wave arrival time(2R(f_(b_cfar))/C_(o)) from R(f_(b_cfar)) are different.

That is, when the output of the first Doppler analyzer 210 is used as aphase reference, the phase rotation amount for the noc-th Doppleranalyzer 210 is given by 2π(noc−1)Δf×(2R(f_(b_cfar))/C_(o)). Tocompensate for the phase rotation, in equation 81-12, the term

$\begin{matrix}\lbrack 95\rbrack &  \\{\exp\lbrack {{- j}2{\pi( {{noc} - 1} )}\Delta f\frac{2{R( f_{b\_ cfar} )}}{C_{0}}} \rbrack} & \end{matrix}$

is derived. Here, noc=1, . . . , Loc.

In this manner, when center frequencies fc of chirp signals areperiodically changed over a plurality of chirp transmission periodswithin code transmission period (Loc×Tr) (or transmission periods, thenumber of which is equal to a divisor of Loc), the period for switchingDoppler analyzers 210 for each code element matches the timing at whichcenter frequencies fc of chirp signals are changed in a round. Forexample, the period in which center frequencies fc of chirp signals areperiodically changed over a plurality of chirp transmission periodsmatches the period for switching Doppler analyzers 210 for each codeelement, thereby allowing coded Doppler demultiplexer 212 d to uniquelydecide a phase correction value for each of Loc Doppler analyzers 210.This enables coded Doppler demultiplexer 212 d to easily perform phasecorrection in (1) code separation process. Therefore, coded Dopplerdemultiplexer 212 d can easily perform the code separation process.

Next, an example of the operation of direction estimator 213 d that isdifferent from that in Embodiment 1 (for example, a conversion processfor Doppler velocity information of the target) will be described.

For example, direction estimator 213 d may calculate Doppler velocityinformation v_(d)(f_(TARGET)) of the target detected based on Dopplerfrequency f_(TARGET) of the target determined by coded Dopplerdemultiplexer 212 d in accordance with following equation 81-13 andoutput Doppler velocity information v_(d)(f_(TARGET)) as a positionmeasurement result.

$\begin{matrix}\lbrack 96\rbrack &  \\{{v_{d}( f_{TARGET} )} = {\frac{C_{0}}{2f_{0}} \times f_{TARGET}}} & ( {{Equation}81 - 13} )\end{matrix}$

As described above, in Variation 10, in radar apparatus 10 d, centerfrequencies fc of chirp signals are periodically changed over aplurality of transmission periods. Even in this case, as in Embodiment1, radar apparatus 10 d (for example, MIMO radar) can extend theeffective Doppler frequency bandwidth to 1/(Tr) and can extend theDoppler frequency (relative velocity) detection range with no ambiguity.In addition, radar apparatus 10 d can reduce mutual interference betweencode-multiplexed signals to about the noise level. According toVariation 10, therefore, radar apparatus 10 d can improve target-objectsensing accuracy over a wider Doppler frequency range.

Variation 10 is applicable not only to Embodiment 1 but also toVariations 1 to 7 of Embodiment 1, and respective similar effects can beachieved. For example, in any of Variations 1 to 7 of Embodiment 1,radar transmission signals including chirp signals whose centerfrequencies fc are changed in the manner illustrated in FIG. 23 may beused.

According to Variation 10, furthermore, the timing at which centerfrequencies fc of chirp signals are changed in a round matches codetransmission period (Loc×Tr). Accordingly, for example, the timing atwhich center frequencies fc of chirp signals are changed in a round canbe made to match the period for switching Doppler analyzers 210 for eachcode element (in other words, the transmission period of a code used forcode multiplexing). This enables coded Doppler demultiplexer 212 d toeasily perform phase correction.

While Variation 10 has described the case where the timing at whichcenter frequencies fc of chirp signals are changed in a round matchescode transmission period (Loc×Tr), the timing at which centerfrequencies fc of chirp signals are changed in a round may be equal to(divisor of (Loc×Tr)). When the timing at which center frequencies fc ofchirp signals are changed in a round is a period given by (divisor E ofLoc×Tr), that is, ε transmission periods (ε×Tr), coded Dopplerdemultiplexer 212 d may use equation 81-14 instead of equation 81-12,where E denotes a divisor of Loc and satisfies ε>1.

$\begin{matrix}{\lbrack 97\rbrack} &  \\{{{DeMUL}_{z}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{noc} = 1}^{Loc}\lbrack {O{C_{ncm}^{*}({noc})}{{VFT}_{z}^{noc}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {{- j}\frac{2{\pi\begin{pmatrix}{f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times}} \\{{\Delta{FD}} - {{DO}P_{1}}}\end{pmatrix}}}{N_{code}} \times \frac{{noc} - 1}{Loc}} \}\exp\{ {{- j}2\pi{mod}( {{{noc} - 1},\varepsilon} )\Delta f\frac{2{R( f_{b\_ cfar} )}}{C_{0}}} \}} \rbrack}} & ( {{Equation}81 - 14} )\end{matrix}$

Further, Variation 10 has described the case where radar transmissionsignals are used for which center frequencies fc of chirp signals arechanged by an integer multiple of Δf, such as f₀, f₀+Δf, f₀+(Loc−1)Δffor each transmission period Tr, in code transmission period (Loc×Tr).However, the change width of center frequencies fc is not limited to afrequency of an integer multiple of Δf, and center frequencies fc may beset to vary by any frequency.

For example, radar transmission signals may be used for which centerfrequencies fc of chirp signals are changed by f₀, f₀+Δf₁, f₀+Δf₂, . . ., and f₀+Δf_(Loc-1) for every transmission period Tr in codetransmission period (Loc×Tr). Here, Δf₁, Δf₂, . . . , and Δf_(Loc-1) arevariable frequency values of center frequencies fc of chirp signals inrespective transmission periods Tr in code transmission period (Loc×Tr).In this case, coded Doppler demultiplexer 212 d may use followingequation 81-15 instead of equation 81-12 in the code separation process.

$\begin{matrix}{\lbrack 98\rbrack} & \\{{{DeMUL}_{z}^{ncm}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{noc} = 1}^{Loc}\lbrack {{{OC}_{ncm}^{*}({noc})}{{VFT}_{z}^{noc}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP_{ndm1}},{DOPCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {{- j}\frac{2{\pi\begin{pmatrix}{f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times}} \\{{\Delta{FD}} - {{DO}P_{1}}}\end{pmatrix}}}{N_{code}} \times \frac{{noc} - 1}{Loc}} \}\exp\{ {{- j}2{\pi\Delta}f_{{noc} - 1}\frac{2{R( f_{b\_ cfar} )}}{C_{0}}} \}} \rbrack}} & ( {{Equation}81 - 15} )\end{matrix}$

Embodiment 2

Embodiment 1 has described the case where both Doppler multiplexingtransmission and coding are used. In contrast, this embodiment describesa case where both Doppler multiplexing transmission and timemultiplexing are used.

In this embodiment, for example, when Doppler-multiplexed transmissionsignals are transmitted in a multiplexed manner also using timemultiplexing (hereinafter referred to as “time-Doppler multiplexing”),the numbers of time multiplexes for the respective Doppler-multiplexedtransmission signals are set to be non-uniform. This enables radarapparatus to identify transmit antennas and determine the presence orabsence of Doppler aliasing based on reception signals of time-Dopplermultiplexed signals. As a result, the effective Doppler frequencybandwidth can be extended to 1/(Tr), and the Doppler frequency (relativevelocity) detection range with no ambiguity can be extended.

[Configuration of Radar Apparatus]

FIG. 24 is a block diagram illustrating an example configuration ofradar apparatus 20 according to this embodiment. In FIG. 24 , componentssimilar to those in Embodiment 1 (FIG. 1 ) are identified with the samenumerals, and a description thereof is omitted.

The following describes, as an example, a configuration of a radarscheme (also referred to as, for example, chirp pulse transmission (fastchirp modulation)) that uses frequency-modulated pulse waves such aschirp pulses. Note that the modulation scheme is not limited to that forfrequency modulation. For example, an exemplary embodiment of thepresent disclosure is also applicable to a radar scheme that uses apulse compression radar configured to transmit a pulse train afterperforming phase modulation or amplitude modulation.

Radar apparatus 20 includes radar transmitter (transmitting branch) 300and radar receiver (receiving branch) 400.

[Configuration of Radar Transmitter 300]

Radar transmitter 300 includes radar transmission signal generator 101,phase rotation amount setter 301, phase rotators 304, transmissioncontrollers 305, and transmit antennas 306.

Phase rotation amount setter 301 sets phase rotation amounts (in otherwords, phase rotation amounts used for Doppler multiplexing) for phaserotators 304. Phase rotation amount setter 301 includes, for example,Doppler shift setter 302 and time multiplexer 303.

For example, Doppler shift setter 302 sets phase rotation amountsϕ_(ndm) corresponding to Doppler shift amounts DOP_(ndm) to be appliedto radar transmission signals (for example, chirp signals) and outputsphase rotation amounts ϕ_(ndm) to time multiplexer 303. Here, ndm=1, . .. , N_(DM). N_(DM) denotes the number of Doppler multiplexes (in otherwords, different numbers of Doppler shifts).

In Radar apparatus 20, since time multiplexing performed by timemultiplexer 303 is also used, the number of Doppler multiplexes N_(DM)may be set to be smaller than the number Nt of transmit antennas 306used for multiplexing transmission. The number of Doppler multiplexesN_(DM) is greater than or equal to, for example, 2.

Doppler shift amounts DOP₁, DOP₂, . . . , and DOP_(DM) are assigneddifferent phase rotation amounts by, for example, dividing a phaserotation range greater than or equal to 0 and less than 2π. For example,phase rotation amount ϕ_(ndm) for applying Doppler shift amountDOP_(ndm) is assigned, as given by equation 1 (the angle is expressed inradian).

The assignment of phase rotation amounts for applying Doppler shiftamounts DOP₁, DOP₂, . . . , and DOP_(DM) is not limited to that in thisassignment method. For example, the assignment of phase rotation amountsgiven by equation 1 may be shifted. For example, phase rotation amountsmay be assigned such that ϕ_(ndm)=2π(ndm)/N_(DM). Alternatively, phaserotation amounts ϕ₁, ϕ₂, . . . , and ϕ_(DM) may be randomly assigned toDoppler shift amounts DOP₁, DOP₂, . . . , and DOP_(DM) using aphase-rotation-amount assignment table.

Time multiplexer 303 sets, for phase rotation amounts ϕ₁, . . . , andϕ_(NDM) for applying N_(DM) Doppler shift amounts output from Dopplershift setter 302, “time-multiplexed Doppler phase rotation amounts”based on the number of time multiplexes N_(TD), and outputstime-multiplexed Doppler phase rotation amounts” to phase rotators 304.

In time multiplexer 303, the number of time multiplexes (hereinafterreferred to as the number of time-Doppler multiplexes) fortime-multiplexing a Doppler multiplexed signal using the ndm-th Dopplershift amount DOP_(ndm) output from Doppler shift setter 302 isrepresented by “N_(DOP_TD)(ndm)”. Here, ndm=1, . . . , N_(DM).

The following describes an example of the operation of time multiplexer303.

Time multiplexer 303 sets the number of time-Doppler multiplexesN_(DOP_TD)(ndm) so that, for example, the sum of the numbers oftime-Doppler multiplexes N_(DOP_TD)(1), N_(DOP_TD)(2), . . . , andN_(DOP_TD)(N_(DM)) to be used to time-multiplex the respective Dopplermultiplexed signals is equal to the number Nt of transmit antennas 306used for multiplexing transmission. In other words, time multiplexer 303sets the number of time-Doppler multiplexes N_(DOP_TD)(ndm) so as tosatisfy following equation 82. This enables radar apparatus 20 toperform multiplexing transmission in the Doppler frequency domain andthe time domain (hereinafter referred to as time-Doppler multiplexingtransmission) using Nt transmit antennas 306.

$\begin{matrix}{{\sum\limits_{{ndm} = 1}^{N_{DM}}{N_{DOP\_ TD}({ndm})}} = {Nt}} & ( {{Equation}82} )\end{matrix}$

Here, time multiplexer 303 sets, for example, the numbers oftime-Doppler multiplexes N_(DOP_TD)(1), N_(DOP_TD)(2), . . . , andN_(DOP_TD)(NDM) so as to include different numbers of time-Dopplermultiplexes in the range greater than or equal to 1 and less than orequal to N_(TD). For example, time multiplexer 303 sets the numbers oftime-Doppler multiplexes such that not all of the numbers oftime-Doppler multiplexes are set to the number of time multiplexesN_(TD), but at least one of the numbers of time-Doppler multiplexes isset to 1 (in other words, no multiplexing). In other words, timemultiplexer 303 sets the numbers of time-Doppler multiplexes for Dopplermultiplexed signals to be non-uniform. With this setting, for example,radar apparatus 20 can individually separate and receive signalstransmitted from plural transmit antennas 306 in a time-Dopplermultiplexed manner through reception processing described below.

Time multiplexer 303 sets, in the m-th transmission period Tr,time-multiplexed Doppler phase rotation amount ψ_(ndop_td(ndm), ndm)(m)given by following equation 83 for phase rotation amount ϕ_(ndm) forapplying the ndm-th Doppler shift amount DOP_(ndm) and outputstime-multiplexed Doppler phase rotation amount ψ_(ndop_td(ndm), ndm)(m)to phase rotators 304.

$\begin{matrix}{{\psi_{{{ndop\_ td}{({ndm})}},{ndm}}(m)} = {{{floor}\lbrack \frac{( {m - 1} )}{N_{TD}} \rbrack} \times \phi_{ndm}}} & ( {{Equation}83} )\end{matrix}$

Here, the subscript “ndop_td(ndm)” represents an index less than orequal to the number of time-Doppler multiplexes N_(DOP_TD)(ndm) forphase rotation amount ϕ_(ndm) for applying Doppler shift amountDOP_(ndm). For example, ndop_td(ndm)=1, . . . , N_(DOP_TD)(ndm).

For example, as given by equation 83, time-multiplexed Doppler phaserotation amount ψ_(ndop_td(ndm), ndm)(m) provides a constant phaserotation amount for applying Doppler shift amount DOP_(ndm) for theduration of N_(TD) transmission periods, the number of which is equal tothe number of time multiplexes used for time multiplexing (for example,N_(TD)×Tr).

Further, time multiplexer 303 outputs, for each transmission period(Tr), time multiplexing index TD_INDEX to radar receiver 400 (outputswitching section 401 described below). Further, time multiplexer 303outputs, for each transmission period Tr, time multiplexing indexTD_INDEX and the number of time-Doppler multiplexes N_(DOP_TD)(ndm) totransmission controllers 305.

TD_INDEX represents a time multiplexing index that indicates atransmission period in the duration of N_(TD) transmission periods(N_(TD)×Tr), the number of which is equal to the number of timemultiplexes, in other words, indicates a transmission duration ortransmission timing for time multiplexing. TD_INDEX cyclically varies inthe range of 1 to N_(TD) for each transmission period (Tr), as given byfollowing equation 84.

TD_INDEX=mod(m−1,N _(TD))+1  (Equation 84)

Here, mod(x, y) denotes a modulo operator and is a function that outputsthe remainder of x divided by y. Further, m=1, . . . , Nc. Nc denotesthe number of transmission periods used for radar position determination(the number of transmissions of radar transmission signals). The numberNc of transmissions of radar transmission signals is set to an integermultiple (Ntdslot times) of N_(TD). For example, Nc=N_(TD)×Ntdslot.

Next, an example method for setting the numbers of time-Dopplermultiplexes N_(DOP_TD)(ndm) for Doppler multiplexed signals to benon-uniform using time multiplexer 303 will be described.

For example, time multiplexer 303 sets the number of time multiplexesN_(TD) satisfying the condition below. For example, the number of timemultiplexes N_(TD) and the number of Doppler multiplexes N_(DM) satisfythe following relationship for the number Nt of transmit antennas 306used for multiplexing transmission.

(Number of time multiplexes N _(TD))×(number of Doppler multiplexes N_(DM))>number Nt of transmit antennas used for multiplexing transmission

For example, among the numbers of time multiplexes N_(TD) and thenumbers of Doppler multiplexes N_(DM) satisfying the above-describedcondition, the use of a combination having a smaller value of theproduct (N_(TD)×N_(DM)) is desirable in terms of both characteristicsand complexity of circuit configuration. Note that among the numbers oftime multiplexes N_(TD) and the numbers of Doppler multiplexes N_(DM)satisfying the above-described condition, a combination having a smallervalue of the product (N_(TD)×N_(DM)) is not restrictive, and any othercombination may be applied.

For example, in a case where Nt=3, the combination of N_(DM)=2, N_(TD)=2is desirable.

In this case, in the assignment of Doppler shift amounts DOP₁ and DOP₂and time multiplexes, the numbers of time-Doppler multiplexesN_(DOP_TD)(1) and N_(DOP_TD)(2) are set in two combinations, namely, thecombination of N_(DOP_TD)(1)=2 and N_(DOP_TD)(2)=1, and the combinationof N_(DOP_TD)(1)=1 and N_(DOP_TD)(2)=2.

Further, for example, in a case where Nt=4, the combination of N_(DM)=3and N_(TD)=2 or the combination of N_(DM)=2 and N_(TD)=3 is desirable.

For example, in a case where Nt=4, N_(DM)=3, and N_(TD)=2, in theassignment of Doppler shift amounts DOP₁, DOP₂, and DOP₃ and timemultiplexes, the numbers of time-Doppler multiplexes N_(DOP_TD)(1),N_(DOP_TD)(2), and N_(DOP_TD)(3) are set in three combinations, namely,the combination of N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=1, andN_(DOP_TD)(3)=1, the combination of N_(DOP_TD)(1)=1, N_(DOP_TD)(2)=2,and N_(DOP_TD)(3)=1, and the combination of N_(DOP_TD)(1)=1,N_(DOP_TD)(2)=1, and N_(DOP_TD)(3)=2.

For example, in a case where Nt=4, N_(DM)=2, and N_(TD)=3, in theassignment of Doppler shift amounts DOP₁ and DOP₂ and time multiplexes,the numbers of time-Doppler multiplexes N_(DOP_TD)(1) and N_(DOP_TD)(2)are set in two combinations, namely, the combination of N_(DOP_TD)(1)=3and N_(DOP_TD)(2)=1, and the combination of N_(DOP_TD)(1)=1 andN_(DOP_TD)(2)=3.

Further, for example, in a case where Nt=5, the combination of N_(DM)=3and N_(TD)=2 is desirable.

For example, in a case where Nt=5, N_(DM)=3, and N_(TD)=2, in theassignment of Doppler shift amounts DOP₁, DOP₂, and DOP₃ and timemultiplexes, the numbers of time-Doppler multiplexes N_(DOP_TD)(1),N_(DOP_TD)(2), and N_(DOP_TD)(3) are set in three combinations, namely,the combination of N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=2, andN_(DOP_TD)(3)=1, the combination of N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=1,and N_(DOP_TD)(3)=2, and the combination of N_(DOP_TD)(1)=1,N_(DOP_TD)(2)=2, and N_(DOP_TD)(3)=2.

Further, for example, in a case where Nt=6 or 7, the combination ofN_(DM)=4 and N_(TD)=2 is desirable.

For example, in a case where Nt=6, N_(DM)=4, and N_(TD)=2, in theassignment of Doppler shift amounts DOP₁, DOP₂, DOP₃, and DOP₄ and timemultiplexes, the numbers of time-Doppler multiplexes N_(DOP_TD)(1),N_(DOP_TD)(2), N_(DOP_TD)(3), and N_(DOP_TD)(4) are set in following sixcombinations.

-   -   N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=2, N_(DOP_TD)(3)=1,        N_(DOP_TD)(4)=1    -   N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=1, N_(DOP_TD)(3)=2,        N_(DOP_TD)(4)=1    -   N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=1, N_(DOP_TD)(3)=1,        N_(DOP_TD)(4)=2    -   N_(DOP_TD)(1)=1, N_(DOP_TD)(2)=2, N_(DOP_TD)(3)=2,        N_(DOP_TD)(4)=1    -   N_(DOP_TD)(1)=1, N_(DOP_TD)(2)=2, N_(DOP_TD)(3)=1,        N_(DOP_TD)(4)=2    -   N_(DOP_TD)(1)=1, N_(DOP_TD)(2)=1, N_(DOP_TD)(3)=2,        N_(DOP_TD)(4)=2

For example, in a case where Nt=7, N_(DM)=4, and N_(TD)=2, in theassignment of Doppler shift amounts DOP₁, DOP₂, DOP₃, and DOP₄ and timemultiplexes, the numbers of time-Doppler multiplexes N_(DOP_TD)(1),N_(DOP_TD)(2), N_(DOP_TD)(3), and N_(DOP_TD)(4) are set in followingfour combinations.

-   -   N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=2, N_(DOP_TD)(3)=2,        N_(DOP_TD)(4)=1    -   N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=2, N_(DOP_TD)(3)=1,        N_(DOP_TD)(4)=2    -   N_(DOP_TD)(1)=2, N_(DOP_TD)(2)=1, N_(DOP_TD)(3)=2,        N_(DOP_TD)(4)=2    -   N_(DOP_TD)(1)=1, N_(DOP_TD)(2)=2, N_(DOP_TD)(3)=2,        N_(DOP_TD)(4)=1

Likewise, for example, in a case where Nt=8 or 9, the combination ofN_(DM)=5 and N_(TD)=2 is desirable. Further, for example, in a casewhere Nt=10, the combination of N_(DM)=6 and N_(TD)=2 is desirable. Thenumber Nt of transmit antennas 306 is not limited to that in the exampledescribed above, and an exemplary embodiment of the present disclosureis also applicable to Nt=11 or more.

Next, an example of how time-multiplexed Doppler phase rotation amountψ_(ndop_td(ndm), ndm)(m) is set will be described.

For example, a description will be given of a case where in timemultiplexer 303, the number of transmit antennas used for multiplexingtransmission Nt=3, the number of Doppler multiplexes N_(DM)=2, andN_(TD)=2. In this case, for example, if the numbers of time-Dopplermultiplexes are set such that N_(DOP_TD)(1)=1 and N_(DOP_TD) (2)=2, timemultiplexer 303 sets time-multiplexed Doppler phase rotation amountsψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) given by following equations85 to 87 and outputs time-multiplexed Doppler phase rotation amountsψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) to phase rotators 304. Here,m=1, . . . , Nc.

[102]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }={0,0,ϕ₁,ϕ₁,2ϕ₁,2ϕ₁,3ϕ₁,3ϕ₁, . . . }  (Equation 85)

[103]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }={0,0,ϕ₂,ϕ₂,2ϕ₂,2ϕ₂,3ϕ₂,3ϕ₂, . . . }  (Equation 86)

[104]

{ψ_(2,2)(1),ψ_(2,2)(2),ψ_(2,2)(3),ψ_(2,2)(4),ψ_(2,2)(5),ψ_(2,2)(6),ψ_(2,2)(7),ψ_(2,2)(8),. . . }={0,0,ϕ₂,ϕ₂,2ϕ₂,2ϕ₂,3ϕ₂,3ϕ₂, . . . }  (Equation 87)

In equations 86 and 87, time-multiplexed Doppler phase rotation amountsψ_(1, 2)(m) and ψ_(2, 2)(m) are the same phase rotation amount, and thecorresponding signals are subjected to, for example, an operation oftemporally shifting transmission timings (time multiplexing) intransmission controllers 305 described below.

Here, as an example, a description will be given of a case where thephase rotation amount for applying Doppler shift amount DOP_(ndm) isgiven by ϕ_(ndm)=2π(ndm−1)/N_(DM) in equation 1, and phase rotationamount ϕ₁ for applying Doppler shift amount DOP₁, which is equal to 0,and phase rotation amount ϕ₂ for applying Doppler shift amount DOP₂,which is equal to π, are used. That is, intervals of phase rotationamounts ϕ_(ndm) for applying Doppler shift amounts DOP_(ndm) are equal.In this case, time multiplexer 303 sets coded Doppler phase rotationamounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) given by followingequations 88 to 90 and outputs coded Doppler phase rotation amountsψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) to phase rotators 304. Here,m=1, . . . , Nc. Here, a modulo operation for 2π is performed, andresults are expressed in radians ranging from 0 or more to less than 2π(the same applies to the following description).

[105]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }={0,0,0,0,0,0,0,0, . . . }  (Equation 88)

[106]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }={0,0,π,π,0,0,π,π, . . . }  (Equation 89)

[107]

{ψ_(2,2)(1),ψ_(2,2)(2),ψ_(2,2)(3),ψ_(2,2)(4),ψ_(2,2)(5),ψ_(2,2)(6),ψ_(2,2)(7),ψ_(2,2)(8),. . . }={0,0,π,π,0,0,π,π, . . . }  (Equation 90)

As given by equations 88 to 90, when a phase rotation amount is set toϕ_(ndm)=2π(ndm−1)/N_(DM), into which 2π is equally divided,time-multiplexed Doppler phase rotation amounts ψ_(1, 1)(m),ψ_(1, 2)(m), and ψ_(2, 2)(m) are changed in transmission periods givenby N_(DM)×N_(TD)=2×2=4.

As given by equations 88 to 90, furthermore, the number of phases usedfor a phase rotation amount for applying a Doppler shift amount (forexample, two, namely, 0 and π) is equal to the number of Doppler shiftamounts used for multiplexing transmission (in other words, the numberof Doppler multiplexes) N_(DM)=2.

Further, for example, a description will be given of a case where intime multiplexer 303, the number of transmit antennas used formultiplexing transmission Nt=6, the number of Doppler multiplexesN_(OM)=4, and N_(TD)=2. In this case, for example, if the numbers oftime-Doppler multiplexes are set such that N_(DOP_TD) (1)=1, N_(DOP_TD)(2)=1, N_(DOP_TD) (3)=2, and N_(DOP_TD) (4)=2, time multiplexer 303 setstime-multiplexed Doppler phase rotation amounts ψ_(1, 1)(m),ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m)given by following equations 91 to 96 and outputs time-multiplexedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ_(1, 3)(m),ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m) to phase rotators 304. Here,m=1, . . . , Nc.

[108]

{ψ_(1,1)(1),ψ_(1,1)(2),ψ_(1,1)(3),ψ_(1,1)(4),ψ_(1,1)(5),ψ_(1,1)(6),ψ_(1,1)(7),ψ_(1,1)(8),. . . }={0,0,ϕ₁,ϕ₁,2ϕ₁,2ϕ₁,3ϕ₁,3ϕ₁, . . . }  (Equation 91)

[109]

{ψ_(1,2)(1),ψ_(1,2)(2),ψ_(1,2)(3),ψ_(1,2)(4),ψ_(1,2)(5),ψ_(1,2)(6),ψ_(1,2)(7),ψ_(1,2)(8),. . . }={0,0,ϕ₂,ϕ₂,2ϕ₂,2ϕ₂,3ϕ₂,3ϕ₂, . . . }  (Equation 92)=

[110]

{ψ_(1,3)(1),ψ_(1,3)(2),ψ_(1,3)(3),ψ_(1,3)(4),ψ_(1,3)(5),ψ_(1,3)(6),ψ_(1,3)(7),ψ_(1,3)(8),. . . }={0,0,ϕ₃,ϕ₃,2ϕ₃,2ϕ₃,3ϕ₃,3ϕ₃, . . . }  (Equation 93)

[111]

{ψ_(2,3)(1),ψ_(2,3)(2),ψ_(2,3)(3),ψ_(2,3)(4),ψ_(2,3)(5),ψ_(2,3)(6),ψ_(2,3)(7),ψ_(2,3)(8),. . . }={0,0,ϕ₃,ϕ₃,2ϕ₃,2ϕ₃,3ϕ₃,3ϕ₃, . . . }  (Equation 94)

[112]

{ψ_(1,4)(1),ψ_(1,4)(2),ψ_(1,4)(3),ψ_(1,4)(4),ψ_(1,4)(5),ψ_(1,4)(6),ψ_(1,4)(7),ψ_(1,4)(8),. . . }={0,0,ϕ₄,ϕ₄,2ϕ₄,2ϕ₄,3ϕ₄,3ϕ₄, . . . }  (Equation 95)

[113]

{ψ_(2,4)(1),ψ_(2,4)(2),ψ_(2,4)(3),ψ_(2,4)(4),ψ_(2,4)(5),ψ_(2,4)(6),ψ_(2,4)(7),ψ_(2,4)(8),. . . }={0,0,ϕ₄,ϕ₄,2ϕ₄,2ϕ₄,3ϕ₄,3ϕ₄, . . . }  (Equation 96)

In equations 93 and 94, time-multiplexed Doppler phase rotation amountsψ_(1, 3)(m) and ψ_(2, 3)(m) are the same phase rotation amount, and thecorresponding signals are subjected to an operation of temporallyshifting transmission timings (time multiplexing) in transmissioncontrollers 305 described below. Also in equations 95 and 96,time-multiplexed Doppler phase rotation amounts ψ_(1, 4)(m) andψ_(2, 4)(m) are the same phase rotation amount, and the correspondingsignals are subjected to an operation of temporally shiftingtransmission timings (time multiplexing) in transmission controllers 305described below.

Here, as an example, the phase rotation amount for applying Dopplershift amount DOP_(ndm) is given by ϕ_(ndm)=2π(ndm−1)/N_(DM), and phaserotation amount ϕ₁ for applying Doppler shift amount DOP₁, which isequal to 0, phase rotation amount ϕ₂ for applying Doppler shift amountDOP₂, which is equal to π/2, phase rotation amount ϕ₃ for applyingDoppler shift amount DOP₃, which is equal to π, and phase rotationamount ϕ₄ for applying Doppler shift amount DOP₄, which is equal to3π/2, are used. In this case, time multiplexer 303 sets time-multiplexedDoppler phase rotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), ψ_(1, 3)(m),ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m) given by following equations97 to 102 and outputs time-multiplexed Doppler phase rotation amountsψ_(1, 1)(m), ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m), ψ_(1, 4)(m), andψ_(2, 4)(m) to phase rotators 304. Here, m=1, . . . , Nc.

$\begin{matrix}{\{ {{\psi_{1,1}(1)},{\psi_{1,1}(2)},{\psi_{1,1}(3)},{\psi_{1,1}(4)},{\psi_{1,1}(5)},{\psi_{1,1}(6)},{\psi_{1,1}(7)},{\psi_{1,1}(8)},\ldots} \} = \{ {0,0,0,0,0,0,0,0,\ldots} \}} & ( {{Equation}97} )\end{matrix}$ $\begin{matrix}{\{ {{\psi_{1,2}(1)},{\psi_{1,2}(2)},{\psi_{1,2}(3)},{\psi_{1,2}(4)},{\psi_{1,2}(5)},{\psi_{1,2}(6)},{\psi_{1,2}(7)},{\psi_{1,2}(8)},\ldots} \} = \{ {0,0,\frac{\pi}{2},\frac{\pi}{2},\pi,\pi,\frac{3\pi}{2},\frac{3\pi}{2},\ldots} \}} & ( {{Equation}98} )\end{matrix}$ $\begin{matrix}{\{ {{\psi_{1,3}(1)},{\psi_{1,3}(2)},{\psi_{1,3}(3)},{\psi_{1,3}(4)},{\psi_{1,3}(5)},{\psi_{1,3}(6)},{\psi_{1,3}(7)},{\psi_{1,3}(8)},\ldots} \} = \{ {0,0,\pi,\pi,0,0,\pi,\pi,\ldots} \}} & ( {{Equation}99} )\end{matrix}$ $\begin{matrix}{\{ {{\psi_{2,3}(1)},{\psi_{2,3}(2)},{\psi_{2,3}(3)},{\psi_{2,3}(4)},{\psi_{2,3}(5)},{\psi_{2,3}(6)},{\psi_{2,3}(7)},{\psi_{2,3}(8)},\ldots} \} = \{ {0,0,\pi,\pi,0,0,\pi,\pi,\ldots} \}} & ( {{Equation}100} )\end{matrix}$ $\begin{matrix}{\{ {{\psi_{1,4}(1)},{\psi_{1,4}(2)},{\psi_{1,4}(3)},{\psi_{1,4}(4)},{\psi_{1,4}(5)},{\psi_{1,4}(6)},{\psi_{1,4}(7)},{\psi_{1,4}(8)},\ldots} \} = \{ {0,0,\frac{3\pi}{2},\frac{3\pi}{2},\pi,\pi,\frac{\pi}{2},\frac{\pi}{2},\ldots} \}} & ( {{Equation}101} )\end{matrix}$ $\begin{matrix}{\{ {{\psi_{2,4}(1)},{\psi_{2,4}(2)},{\psi_{2,4}(3)},{\psi_{2,4}(4)},{\psi_{2,4}(5)},{\psi_{2,4}(6)},{\psi_{2,4}(7)},{\psi_{2,4}(8)},\ldots} \} = \{ {0,0,\frac{3\pi}{2},\frac{3\pi}{2},\pi,\pi,\frac{\pi}{2},\frac{\pi}{2},\ldots} \}} & ( {{Equation}102} )\end{matrix}$

As given by equations 97 to 102, when a phase rotation amount is set toϕ_(ndm)=2π(ndm−1)/N_(DM), into which 2π is equally divided,time-multiplexed Doppler phase rotation amounts ψ_(1, 1)(m),ψ_(1, 2)(m), ψ_(1, 3)(m), ψ_(2, 3)(m), ψ_(1, 4)(m), and ψ_(2, 4)(m) arechanged in transmission periods given by N_(DM)×N_(TD)=4×2=8.

As given by equations 97 to 102, furthermore, the number of phases usedfor a phase rotation amount for applying a Doppler shift amount (forexample, four, namely, 0, π/2, π, and 3π/2) is equal to the number ofDoppler shift amounts used for multiplexing transmission (in otherwords, the number of Doppler multiplexes) N_(DM)=4.

While the description has been given of, as an example, the setting ofphase rotation amounts in a case where the number Nt of transmitantennas 306 is equal to 3 and the number of Doppler multiplexes N_(DM)is equal to 2 and in a case where the number Nt of transmit antennas 306is equal to 6 and the number of Doppler multiplexes N_(DM) is equal to4, the number Nt of transmit antennas 306 and the number of Dopplermultiplexes N_(DM) are not limited to the values described above. Forexample, the number of phases used for a phase rotation amount forapplying a Doppler shift amount may be equal to the number N_(DM) ofDoppler shift amounts used for multiplexing transmission.

The foregoing description has been given of a method for setting phaserotation amounts using time multiplexer 303.

In FIG. 24 , each phase rotator 304 applies a phase rotation amount to achirp signal output from radar transmission signal generator 101, foreach transmission period Tr, based on the time-multiplexed Doppler phaserotation amount ψ_(ndop_td(ndm), ndm)(m) set by phase rotation amountsetter 301. Here, ndm=1, . . . , N_(DM), and ndop_td(ndm)=1, . . . ,N_(DOP_TD)(ndm).

The sum of the numbers of time-Doppler multiplexes N_(DOP_TD)(1),N_(DOP_TD)(2), . . . , and N_(DOP_TD)(N_(DM)) is set to be equal to thenumber Nt of transmit antennas 306, and Nt time-multiplexed Dopplerphase rotation amounts are respectively input to Nt phase rotators 304.

Each of Nt phase rotators 304 applies time-multiplexed Doppler phaserotation amount ψ_(ndop_td(ndm), ndm)(m) input thereto to a chirp signaloutput from radar transmission signal generator 101 for eachtransmission period Tr. In the following, phase rotator 304 that appliestime-multiplexed Doppler phase rotation amount ψ_(ndop_td(ndm), ndm)(m)is represented by “phase rotator PROT #[ndop_td(ndm), ndm]”.

The outputs of Nt phase rotators 304 are output to Nt transmissioncontrollers 305. For example, the output of phase rotator PROT#[ndop_td(ndm), ndm] is output to transmission controller TXCTRL#[ndop_td(ndm), ndm]. Here, ndm=1, . . . , N_(DM), and ndop_td(ndm)=1, .. . , N_(DOP_TD)(ndm).

In FIG. 24 , among Nt transmission controllers 305, transmissioncontroller TXCTRL #[ndop_td(ndm), ndm] controls, based on the number oftime-Doppler multiplexes N_(DOP_TD)(ndm) and time multiplexing indexTD_INDEX, the transmission (in other words, output) of a transmissionsignal input from the corresponding one of phase rotators 304. Forexample, transmission controllers 305 controls the turning on and off oftransmission (transmission on and transmission off) of transmissionsignals.

In the following, whether transmission controller TXCTRL #[ndop_td(ndm),ndm] is in transmission on or transmission off state for each timemultiplexing index TD_INDEX is displayed using “transmission controllerresponse STATE_TXCTRL #[ndop_td(ndm), ndm]=[state(1), state(2), . . . ,state(N_(TD))]”.

An element of STATE_TXCTRL #[ndop_td(ndm), ndm], namely,state(TD_INDEX), indicates a transmission on or transmission off statefor TD_INDEX=1, . . . , N_(TD). For example, state(TD_INDEX) mayindicate 1 for transmission on, and state(TD_INDEX) may indicate 0 fortransmission off.

For example, when the number of time-Doppler multiplexesN_(DOP_TD)(ndm)=1 for a Doppler multiplexed signal using Doppler shiftamount DOP_(ndm), ndop_td(ndm)=1, and the number of time multiplexesis 1. Thus, transmission controller TXCTRL #[1, ndm] outputs alltransmission signals for time multiplexing indices TD_INDEX=1, . . . ,N_(TD) (transmission on). Accordingly, when the number of time-Dopplermultiplexes N_(DOP_TD)(ndm)=1 for a Doppler multiplexed signal usingDoppler shift amount DOP_(ndm), the transmission controller response isrepresented by STATE_TXCTRL #[1, ndm]=[1, 1, . . . , 1].

In contrast, for example, when the number of time-Doppler multiplexesN_(DOP_TD)(ndm)=N_(TD) for a Doppler multiplexed signal using Dopplershift amount DOP_(ndm), ndop_td(ndm)=1, . . . , N_(TD), and the numberof time multiplexes is N_(TD). Thus, transmission controller TXCTRL#[ndop_td(ndm), ndm] outputs a transmission signal when timemultiplexing index TD_INDEX=ndop_td(ndm) (transmission on). Accordingly,when the number of time-Doppler multiplexesN_(DOP_TD)(ndm)=N_(DOP_TD)(ndm)=N_(TD) for a Doppler multiplexed signalusing Doppler shift amount DOP_(ndm), the transmission controllerresponse STATE_TXCTRL #[ndop_td(ndm), ndm] is 1 (transmission on) whenTD_INDEX=ndop_td(ndm), and is 0 (transmission off) whenTD_INDEX=ndop_td(ndm).

For example, when the number of time-Doppler multiplexes N_(DOP_TD)(1)=4(=N_(TD)) for a Doppler multiplexed signal using Doppler shift amountDOP₁, the transmission controller response for transmission controllerTXCTRL #[1, 1] is represented by STATE_TXCTRL #[1, 1]=[1, 0, 0, 0], thetransmission controller response for transmission controller TXCTRL #[2,1] is represented by STATE_TXCTRL #[2, 1]=[0, 1, 0, 0], the transmissioncontroller response for transmission controller TXCTRL #[3, 1] isrepresented by STATE_TXCTRL #[3, 1]=[0, 0, 1, 0], and the transmissioncontroller response for transmission controller TXCTRL #[4, 1] isrepresented by STATE_TXCTRL #[4, 1]=[0, 0, 0, 1].

The outputs of transmission controllers 305 (referred to as, forexample, time-Doppler multiplexed signals) are amplified to definedtransmission power and are then radiated into a space from Nt transmitantennas 306 in a transmit array antenna section. In other words, phaserotation amounts corresponding to Doppler shift amounts are applied toradar transmission signals, which are then time-multiplexed andtransmitted from plural transmit antennas 306.

In the following, transmit antenna 306 that radiates the output oftransmission controller TXCTRL #[ndop_td(ndm), ndm] into a space isrepresented by “transmit antenna Tx #[ndop_td(ndm), ndm]”. Here, ndm=1,. . . , N_(DM), and ndop_code(ndm)=1, . . . , N_(DOP_CODE)(ndm).

For example, in a case where the number of transmit antennas used formultiplexing transmission Nt=3, the number of Doppler multiplexesN_(DM)=2, N_(TD)=2, and numbers of time-Doppler multiplexes are set suchthat N_(DOP_TD)(1)=1 and N_(DOP_TD)(2)=2, time-multiplexed Doppler phaserotation amounts ψ_(1, 1)(m), ψ_(1, 2)(m), and ψ_(2, 2)(m) are outputfrom time multiplexer 303 to phase rotators 304 for every transmissionperiod.

For example, phase rotator PROT #[1, 1] applies, for each transmissionperiod, phase rotation amount ψ_(1, 1)(m) to a chirp signal generated inradar transmission signal generator 101 for each transmission period.The output of phase rotator PROT #[1, 1] is output in a manner given byfollowing expression 103 based on transmission controller responseSTATE_TXCTRL #[1, 1]=[1,1] for transmission controller TXCTRL #[1, 1],and is transmitted from transmit antenna Tx #[1, 1]. Here, cp(t) denotesa chirp signal for each transmission period.

exp[jψ _(1,1)(1)]cp(t),exp[jψ _(1,1)(2)]cp(t),exp[jψ _(1,1)(3)]cp(t), .. . ,exp[jψ _(1,1)(Nc)]cp(t)   (Expression 103)

Likewise, phase rotator PROT #[1, 2] applies, for each transmissionperiod, phase rotation amount ψ_(1, 2)(m) to a chirp signal generated inradar transmission signal generator 101 for each transmission period.The output of phase rotator PROT #[1, 2] is output in a manner given byfollowing expression 104 based on transmission controller responseSTATE_TXCTRL #[1, 2]=[1,0] for transmission controller TXCTRL #[1, 2],and is transmitted from transmit antenna Tx #[1, 2].

exp[jψ _(1,2)(1)]cp(t),exp[jψ _(1,2)(2)]cp(t),exp[jψ _(1,2)(3)]cp(t), .. . ,exp[jψ _(1,2)(Nc)]cp(t)   (Expression 104)

Likewise, phase rotator PROT #[2, 2] applies, for each transmissionperiod, phase rotation amount ψ_(2, 2)(m) to a chirp signal generated inradar transmission signal generator 101 for each transmission period.The output of phase rotator PROT #[2, 2] is output in a manner given byfollowing expression 105 based on transmission controller responseSTATE_TXCTRL #[2, 2]=[0,1] for transmission controller TXCTRL #[2, 2],and is transmitted from transmit antenna Tx #[2, 2].

exp[jψ _(2,2)(1)]cp(t),exp[jψ _(2,2)(2)]cp(t),exp[jψ _(2,2)(3)]cp(t), .. . ,exp[jψ _(2,2)(Nc)]cp(t)   (Expression 105)

The foregoing description has been given of an example of howtime-multiplexed Doppler phase rotation amount ψ_(ndop_td(ndm), ndm)(m)is set.

In this embodiment, accordingly, plural transmit antennas 306 areassociated with combinations (in other words, assignment) of Dopplershift amounts DOP_(ndm) and transmission durations for time multiplexing(for example, TD_INDEX) such that in each of the combinations, at leastone of Doppler shift amount DOP_(ndm) or the transmission duration fortime multiplexing (for example, TD_INDEX) is different. In thisembodiment, furthermore, the number of time multiplexes (in other words,the number of time-Doppler multiplexes N_(DOP_TD)(ndm)) corresponding toeach Doppler shift amount DOP_(ndm) in combinations of Doppler shiftamounts DOP_(ndm) and transmission durations is different.

For example, in this embodiment, Nt transmit antennas 306 include atleast plural transmit antennas 306 from which transmission signals thatare time-multiplexed are transmitted, and at least one transmit antenna306 from which a transmission signal that is not time-multiplexed istransmitted. In other words, radar transmission signals transmitted fromradar transmitter 300 include at least a time-Doppler multiplexed signalfor which the number of time-Doppler multiplexes N_(DOP_TD)(ndm) is setto the number of time multiplexes N_(TD), and a time-Doppler multiplexedsignal for which the number of time-Doppler multiplexes N_(DOP_TD)(ndm)is set to be smaller than the number of codes N_(TD).

[Configuration of Radar Receiver 400]

In FIG. 24 , output switching section 401 in the z-th signal processor206 selectively switches and outputs the output of beat frequencyanalyzer 208 for each transmission period to the TD_INDEX-th Doppleranalyzer 402 among N_(TD) Doppler analyzers 402, based on timemultiplexing index TD_INDEX output from time multiplexer 303 in phaserotation amount setter 301. In other words, output switching section 401selects the TD_INDEX-th Doppler analyzer 402 in the m-th transmissionperiod Tr.

The z-th signal processor 206 includes N_(TD) Doppler analyzers 402-1 to402-N_(TD). For example, data is input to the ntd-th Doppler analyzer402 in the z-th signal processor 206 for every N_(TD) transmissionperiods (N_(TD)×Tr) by using output switching section 401. Accordingly,the ntd-th Doppler analyzer 402 performs Doppler analysis for eachdistance index f_(b) using data of Ntdslot transmission periods among Nctransmission periods. Here, ntd denotes the index for time multiplexing,and ntd=1, . . . , N_(TD).

For example, when Ntdslot is a power of 2, FFT processing is applicablein Doppler analysis. In this case, the FFT size is Ntdslot, and amaximum Doppler frequency without causing aliasing derived from thesampling theorem is ±1/(2N_(TD)×Tr). Further, the Doppler frequencyinterval for Doppler frequency index f_(s) is 1/(Ntdslot×N_(TD)×Tr), andthe range of Doppler frequency index f_(s) is given by f_(s)=−Ntdslot/2,. . . , 0, . . . , Ntdslot/2−1.

The following description will be given of a case where Ntdslot is apower of 2, as an example. When Ntdslot is not a power of 2, forexample, data with zero padding can be used to perform FFT processingwith a number of data sizes (FFT sizes) equal to a power of 2. In theFFT processing, Doppler analyzer 402 may perform multiplication by awindow function coefficient such as of the Han window or the Hammingwindow. The application of a window function can suppress sidelobesaround the beat frequency peak.

For example, output VFT_(z) ^(ntd)(f_(b), f_(s)) of Doppler analyzers402 in the z-th signal processor 206 is given by following equation 106,where j is the imaginary unit and z=1 to Na.

$\begin{matrix}{{{VFT}_{z}^{ntd}( {f_{b},f_{s}} )} = {\sum\limits_{s = 0}^{N_{tdslot} - 1}{{{RFT}_{z}( {f_{b},{{N_{TD} \times s} + {ntd}}} )}{\exp\lbrack {{- j}\frac{2\pi{sf}_{s}}{N_{tdslot}}} \rbrack}}}} & ( {{Equation}106} )\end{matrix}$

In FIG. 24 , CFAR section 403 performs CFAR processing (in other words,adaptive threshold determination) using the outputs of N_(TD) Doppleranalyzers 402 in each of the first to Na-th signal processors 206 andextracts distance index f_(b_cfar) and Doppler frequency indexf_(s_cfar) that provide a peak signal.

For example, CFAR section 403 performs two-dimensional CFAR processingwith the distance axis and the Doppler frequency axis (corresponding tothe relative velocity) or CFAR processing that is a combination ofone-dimensional CFAR processing operations by power addition of outputsVFT_(z) ^(ntd)(f_(b), f_(s)) of Doppler analyzers 402 in the first toNa-th signal processors 206, for example, as given by following equation107. As the two-dimensional CFAR processing or the CFAR processing thatis a combination of one-dimensional CFAR processing operations, forexample, processing disclosed in NPL 2 may be applied.

$\begin{matrix}{{{PowerFT}( {f_{b},f_{s}} )} = {\sum\limits_{z = 1}^{N_{a}}{\sum\limits_{{ntd} = 1}^{N_{TD}}{❘{{VFT}_{z}^{ntd}( {f_{b},\ f_{s}} )}❘}^{2}}}} & ( {{Equation}107} )\end{matrix}$

CFAR section 403 adaptively sets a threshold and outputs distance indexf_(b_cfar) that provides received power greater than the threshold,Doppler frequency index f_(s_cfar), and received power informationPowerFT(f_(b_cfar), f_(s_cfar)) to time-Doppler demultiplexer 404.

For example, when phase rotation amount ϕ_(ndm) for applying Dopplershift amount DOP_(ndm) is determined using equation 1, intervals ΔFD ofDoppler shift amounts in the Doppler frequency domain, which are outputfrom Doppler analyzers 402, are equal, where ΔFD=Ntdslot/N_(DM).Accordingly, in the outputs of Doppler analyzers 402, peaks are detectedat intervals of ΔFD for Doppler-shift multiplexed signals in the Dopplerfrequency domain. When phase rotation amount ϕ_(ndm) is determined usingequation 1, ΔFD may not sometimes be an integer depending on Ntdslot andN_(DM). In this case, following equation 108 may be used to achieve ΔFDhaving an integer value. The following describes a reception processingoperation using ΔFD having an integer value.

$\begin{matrix}{\phi_{ndm} = {\frac{2\pi}{N_{tdslot}}{{round}( \frac{N_{tdslot}}{N_{DM}} )}( {{ndm} - 1} )}} & ( {{Equation}108} )\end{matrix}$

Further, CFAR section 403 may divide the respective outputs of Doppleranalyzers 402 into ranges at Doppler shift amount intervals ΔFD andperform CFAR processing (for example, Doppler domain compression CFARprocessing) after, as given by following equation 109, providing poweraddition (for example, Doppler domain compression) for the respectiveranges while matching peak positions of Doppler-shift multiplexedsignals. Here, f_(s_comp)=−Ntdslot/2, . . . , −Ntdslot/2+ΔFD−1.

$\begin{matrix}{{{PowerFT\_ comp}( {f_{b},\ f_{s\_ comp}} )} = {\sum\limits_{{nfd} = 1}^{N_{DM}}{{PowerFT}( {f_{b},\ {f_{s\_ comp} + {( {{nfd} - 1} ) \times \Delta FD}}} )}}} & ( {{Equation}109} )\end{matrix}$

This can compress the Doppler frequency range for the CFAR processing to1/N_(DM) to reduce the amount of CFAR processing and can simplify thecircuit configuration. In addition, CFAR section 403 enables poweraddition for N_(DM) Doppler-shift multiplexed signals, resulting in SNRbeing improved by about (N_(DM))^(1/2). As a result, the radar sensingperformance of radar apparatus 20 can be improved.

As described above, for example, CFAR section 403, which uses Dopplerdomain compression CFAR processing, adaptively sets a threshold andoutputs distance index f_(b_cfar) that provides received power greaterthan the threshold, Doppler frequency index f_(s_comp_cfar), andreceived power information PowerFT(f_(b_cfar),f_(s_comp_cfar)+(nfd−1)×ΔFD) for Doppler frequency indices(f_(s_comp_cfar)+(nfd−1)×ΔFD) of N_(DM) Doppler multiplexed signals,where nfd=1, . . . , N_(DM), to time-Doppler demultiplexer 404.

Note that phase rotation amount ϕ_(ndm) for applying Doppler shiftamount DOP_(ndm) is not limited to that given in equation 1. CFARsection 403 can apply Doppler domain compression CFAR processing, forexample, if Doppler-shift multiplexed signals have phase rotationamounts ϕ_(ndm) such that peaks are detected at constant intervals inthe Doppler frequency domain output from Doppler analyzers 402.

Next, an example of the operation of time-Doppler demultiplexer 404illustrated in FIG. 24 will be described. The following describes anexample of processing performed by time-Doppler demultiplexer 404 whenCFAR section 403 uses Doppler domain compression CFAR processing.

Time-Doppler demultiplexer 404 separates signals transmitted in atime-Doppler multiplexed manner, using the output of Doppler analyzer402, based on the outputs of CFAR section 403, namely, distance indexf_(b_cfar), Doppler frequency index f_(s_comp_cfar), and received-powerinformation PowerFT(f_(b_cfar), f_(s_comp_cfar)±(nfd−1)×ΔFD) for Dopplerfrequency indices (f_(s_comp_cfar)±(nfd−1)×ΔFD) of N_(DM) Dopplermultiplexed signals, where nfd=1, . . . , N_(DM), and performsdiscrimination of transmit antennas 306 and discrimination of Dopplerfrequencies (or Doppler velocities or relative velocities).

As described above, for example, time multiplexer 303 in phase rotationamount setter 301 does not set all of N_(DM) numbers of time-Dopplermultiplexes N_(DOP_TD)(1), N_(DOP_TD)(2), . . . , and N_(DOP_TD)(N_(DM))to N_(TD), but sets at least one of the numbers of Doppler multiplexesto 1. For example, time-Doppler demultiplexer 404 detects (1) a Dopplermultiplexed signal for which the number of time-Doppler multiplexes isset to 1, and performs (2) discrimination of transmit antennas 306 anddiscrimination of Doppler frequencies of the target based on thedetected Doppler multiplexed signal for which the number of time-Dopplermultiplexes is set to 1.

The following describes the processes (1) and (2) described above, whichare performed by time-Doppler demultiplexer 404.

<(1) Process of Detecting Time-Doppler Multiplexed Signal for whichNumber of Time-Doppler Multiplexes is Set to 1>

There are N_(DM) candidate correspondences between N_(DM) time-Dopplermultiplexed signals and the outputs of the respective Doppler analyzers402 for Doppler frequency indices (f_(s_comp_cfar)),(f_(s_comp_cfar)+ΔFD), (f_(s_comp_cfar)+2ΔFD), . . . , and(f_(s_comp_cfar)+(N_(DM)−1)ΔFD) of N_(DM) time-Doppler multiplexedsignals for distance index f_(b_cfar) output from CFAR section 403.

For example, if Doppler shift amount DOP_(ndm) set in Doppler shiftsetter 302 is represented by DOP₁<DOP₂< . . . <DOP_(DM-1)<DOP_(DM),there are N_(DM) candidate correspondences with cyclically shiftedelements, as follows, with consideration given to Doppler aliasing.Here, patterns of the candidate correspondences are numbered DopCase=1to N_(DM).

-   -   DopCase=1: {DOP₁, DOP₂, . . . , DOP_(DM-1), DOP_(DM)}    -   DopCase=2: {DOP_(DM), DOP₁, DOP₂, . . . , DOP_(DM-1)}    -   . . . ,    -   DopCase=N_(DM): {DOP₂, . . . , DOP_(DM-1), DOP_(DM), DOP₁}

For example, DopCase=1 indicates a correspondence among Doppler shiftamounts in the initial state (when the relative velocity to the targetis zero). For example, more aliasing components are included as therelative velocity of the target increases in a direction in which thedistance to the target decreases, and the resulting correspondences areassociated with DopCase=2, 3, . . . , N_(DM). In other words,DopCase=N_(DM), N_(DM-1), . . . , 2 is associated as the relativevelocity of the target increases in a direction in which the distance tothe target increases.

Here, a table indicating the position of DOP_(ndm) counting from thebeginning of each DopCase (the position (or order) of DOP_(ndm) inDopCase) can be created in advance, based on the Doppler shift amountsset in Doppler shift setter 302. In the following, DOPposi(DOP_(ndm),DopCase) denotes the operator that outputs the position of DOP_(ndm)counting from the beginning of each DopCase. For example, in theabove-described example of DopCase, DOPposi(DOP₁, 1)=1, DOPposi(DOP₁,2)=2, DOPposi(DOP₁, N_(DM))=N_(DM), DOPposi(DOP₂, 1)=2, DOPposi(DOP₂,2)=3, and DOPposi(DOP₂, N_(DM))=1.

Time-Doppler demultiplexer 404 performs a detection process on, forexample, the outputs of Doppler analyzers 402 in the z-th signalprocessor 206 indicated by Doppler frequency indices(f_(s_comp_cfar)+(nfd−1)×ΔFD) of N_(DM) time-Doppler multiplexed signalsfor distance indices f_(b_cfar) output from CFAR section 403 to detectDoppler multiplexed signals for which the numbers of time-Dopplermultiplexes are set to 1.

Here, time-Doppler multiplexed signals for which the numbers oftime-Doppler multiplexes are set to 1 are transmitted using the sametransmit antenna 306, and time-Doppler multiplexed signals for which thenumbers of time-Doppler multiplexes are set to N_(TD) are transmittedusing different transmit antennas 306.

Time-Doppler demultiplexer 404 performs coherent addition processing on,for example, the outputs of Doppler analyzers 402 for each timemultiplexing index TD_INDEX using DopCase as a parameter, withconsideration given to the presence or absence of aliasing. Time-Dopplerdemultiplexer 404 utilizes, for example, DopCase for which a coherentaddition value that realizes maximum power is obtained, as the mostlikely pattern of Doppler shift amounts to identify transmit antennas.

The detection process may be performed on all of nfd=1, . . . , N_(DM)for all the candidates of DopCase=1, . . . , N_(DM). Note that theposition of, in each DopCase, a Doppler multiplexed signal for which thenumber of time-Doppler multiplexes is set to 1 is set by Doppler shiftsetter 302 and is known to radar apparatus 20. Accordingly, time-Dopplerdemultiplexer 404 performs the following detection process to reduce theoperation amount of the separation process.

For example, when a time-Doppler multiplexed signal for which the numberof time Doppler multiplexes is set to 1 is Doppler shift amountDOP_(ndm1), time-Doppler demultiplexer 404 performs coherent additionprocessing on the outputs of Doppler analyzers 402 using candidateDOPposi(DOP_(ndm1), DopCase) including Doppler shift amount DOP_(ndm1)within each DopCase, with consideration given to the presence or absenceof Doppler aliasing. Then, time-Doppler demultiplexer 404 detectsDopCase that realizes maximum power as follows, based on the result ofcoherent addition processing with consideration given to the presence orabsence of Doppler aliasing.

(a) Coherent Addition Processing when Outputs of Doppler Analyzers 402Contain No Doppler Aliasing

Coherent addition signal ADDCOH_(z)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) given by followingequation 110 represents a coherent addition value for the output ofDoppler analyzer 402 for distance index f_(b_cfar) and Doppler frequencyindex (f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) in the z-thsignal processor 206.

$\begin{matrix}{{{ADDCOH}_{z}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{ntd} = 1}^{N_{TD}}\lbrack {{{VFT}_{z}^{ntd}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ \text{ }{{- j}\frac{2{\pi( {f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times \Delta{FD}} - {DOP}_{1}} )}}{N_{tdslot}} \times \frac{{ntd} - 1}{N_{TD}}} \}} \rbrack}} & ( {{Equation}110} )\end{matrix}$

In equation 110, in the exp term, since the sampling times for theoutputs of Doppler analyzers 402 for each time multiplexing index areshifted, phase correction is performed in accordance with Dopplerfrequency index (f_(s_comp_cfar)±(DopCase−1)×ΔFD-DOP₁). In equation 110,furthermore, DopCase=1, . . . , N_(DM). Further, Doppler shift amountDOP_(ndm) set in Doppler shift setter 302 is represented by DOP₁<DOP₂< .. . <DOP_(DM-1)<DOP_(DM), and DOP₁ falls within the range off_(s_comp)=−Ncode/2, . . . , −Ncode/2+ΔFD−1 in the initial state (whenthe relative velocity to the target is zero). Accordingly, time-Dopplerdemultiplexer 404 calculates an amount of phase correction using, forexample, DOP₁ as a reference in equation 110.

When a time-Doppler multiplexed signal for which the number oftime-Doppler multiplexes is set to 1 is Doppler shift amount“DOP_(ndm1)”, for DopCase=1, . . . , N_(DM), coherent addition signalsfor candidate DOPposi(DOP_(ndm1), DopCase) including Doppler shiftamount DOP_(ndm1) are obtained using equation 110. Accordingly,time-Doppler demultiplexer 404 obtains the outputs of a total of N_(DM)coherent addition signals. For example, time-Doppler demultiplexer 404calculates coherent addition signals for all the receive antennas z=1, .. . , Na in accordance with equation 110 and calculates time separationsignal power sum Pow_ADDCOH(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD), as given byfollowing equation 111.

$\begin{matrix}{{{Pow\_ ADDCOH}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \text{ }\Delta{FD}}}} )} = {\sum\limits_{z = 1}^{Na}{❘{{ADDCOH}_{z}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )}❘}^{2}}} & ( {{Equation}111} )\end{matrix}$

(b) Coherent Addition Processing when Outputs of Doppler Analyzers 402Contain Doppler Aliasing

If the outputs of Doppler analyzers 402 contain Doppler aliasing,time-Doppler demultiplexer 404 performs coherent addition processingusing phase correction (the exp term) used in the coherent additionprocessing as phase correction with consideration given to Doppleraliasing.

For example, in a case where N_(TD)=2, coherent addition signalADDALIAS_(z)(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD) given by following equation 112 represents a coherentaddition value with consideration given to Doppler aliasing for theoutput of Doppler analyzer 402 for distance index f_(b_cfar) and Dopplerfrequency index (f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD)in the z-th signal processor 206.

$\begin{matrix}{{{ADDALIAS}_{z}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{ntd} = 1}^{N_{TD}}\lbrack {{{VFT}_{z}^{ntd}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {{- j}\frac{2{\pi( {f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times \Delta{FD}} - {DOP}_{1}} )}}{N_{tdslot}} \times \frac{{ntd} - 1}{N_{TD}}} \} \times \exp\{ {{- j}\frac{2{\pi( {{ntd} - 1} )}}{N_{TD}}} \}} \rbrack}} & ( {{Equation}112} )\end{matrix}$

In equation 112, in the exp term, since the sampling times for theoutputs of Doppler analyzers 402 for each time multiplexing index areshifted, phase correction is performed in accordance with Dopplerfrequency index (f_(s_comp_cfar)±(DopCase−1)×ΔFD-DOP₁) and phasecorrection (exp{−j2π(ntd−1)/N_(TD)}) to address aliasing is performed.In equation 112, furthermore, DopCase=1, . . . , N_(DM). Further,Doppler shift amount DOP_(ndm) set in Doppler shift setter 302 isrepresented by DOP₁<DOP₂< . . . <DOP_(DM-1)<DOP_(DM), and DOP₁ fallswithin the range of fs_comp=−Ncode/2, . . . , −Ncode/2+ΔFD−1 in theinitial state (when the relative velocity to the target is zero).Accordingly, time-Doppler demultiplexer 404 calculates an amount ofphase correction using DOP₁ as a reference in equation 112.

When a time-Doppler multiplexed signal for which the number oftime-Doppler multiplexes is set to 1 is Doppler shift amount DOP_(ndm1),for DopCase=1, . . . , N_(DM), coherent addition signals withconsideration given to Doppler aliasing for candidateDOPposi(DOP_(ndm1), DopCase) including Doppler shift amount DOP_(ndm1)are obtained using Equation 112. Accordingly, time-Doppler demultiplexer404 obtains the outputs of a total of N_(DM) coherent addition signals.For example, time-Doppler demultiplexer 404 calculates coherent additionsignals for all the receive antennas z=1, Na using equation 112 andcalculates coherent addition power sum Pow_ADDALIAS(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) with considerationgiven to Doppler aliasing, as given by following equation 113.

$\begin{matrix}{{{Pow\_ ADDCOH}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{z = 1}^{Na}{❘{{ADDCOH}_{z}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )}❘}^{2}}} & ( {{Equation}113} )\end{matrix}$

For example, in a case where N_(TD)=3, coherent addition signalADDALIAS_(z)(f_(b_cfar), f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD) given by following equations 114 and 115 represents acoherent addition value with consideration given to Doppler aliasing forthe output of Doppler analyzer 402 for distance index f_(b_cfar) andDoppler frequency index (f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD) in the z-th signal processor 206.

$\begin{matrix}{{{ADDALIAS}_{z}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{{ntd} = 1}^{N_{TD}}\lbrack {{{VFT}_{z}^{ntd}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {{- j}\frac{2{\pi( f_{est} )}}{N_{tdslot}} \times \frac{{ntd} - 1}{N_{TD}}} \} \times \exp\{ {{- j}\frac{2\pi{{Sign}( f_{est} )}( {{ntd} - 1} )}{N_{TD}}} \}} \rbrack}} & ( {{Equation}114} )\end{matrix}$ $\begin{matrix}{f_{est} = {f_{{s\_ comp}{\_ cfar}} + {( {{DopCase} - 1} ) \times \Delta{FD}} - {DOP}_{1}}} & ( {{Equation}115} )\end{matrix}$

In equation 114, in the exp term, since the sampling times for theoutputs of Doppler analyzers 402 for each time multiplexing index areshifted, phase correction is performed in accordance with Dopplerfrequency index (f_(s_comp_cfar)±(DopCase−1)×ΔFD−DOP₁) and phasecorrection (exp{−j2πSign(f_(est))(ntd−1)/N_(TD)}) to address aliasing isperformed. In equation 114, furthermore, DopCase=1, . . . , N_(DM).Further, Doppler shift amount DOP_(ndm) set in Doppler shift setter 302is represented by DOP₁<DOP₂< . . . <DOP_(DM-1)<DOP_(DM), and DOP₁ fallswithin the range of f_(s_comp)=−Ncode/2, . . . , −Ncode/2+ΔFD−1 in theinitial state (when the relative velocity to the target is zero).Accordingly, time-Doppler demultiplexer 404 calculates an amount ofphase correction using DOP₁ as a reference in equation 114.

When a time-Doppler multiplexed signal for which the number oftime-Doppler multiplexes is set to 1 is Doppler shift amount DOP_(ndm1),for DopCase=1, . . . , N_(DM), coherent addition signals withconsideration given to Doppler aliasing for candidateDOPposi(DOP_(ndm1), DopCase) including Doppler shift amount DOP_(ndm1)are obtained using Equation 114. Accordingly, time-Doppler demultiplexer404 obtains the outputs of a total of N_(DM) coherent addition signals.For example, time-Doppler demultiplexer 404 calculates coherent additionsignals for all the receive antennas z=1, . . . , Na using equation 114and calculates coherent addition power sum Pow_ADDALIAS(f_(b-efar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) with considerationgiven to Doppler aliasing, as given by following equation 116.

$\begin{matrix}{{{Pow\_ ADDCOH}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )} = {\sum\limits_{z = 1}^{Na}{❘{{ADDCOH}_{z}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + \text{ }{\{ {{{DOPposi}( {{DOP}_{{ndm}1},{DopCase}} )} - 1} \} \times \Delta{FD}}}} )}❘}^{2}}} & ( {{Equation}116} )\end{matrix}$

In a case where N_(TD)=3, coherent addition values with considerationalso given to multiplexed aliasing determination are calculated.

Also in a case where N_(TD)>3, coherent addition valuePow_ADDCOH(f_(b_cfar), f_(s_comp_cfar)±(DOPposi(DOP_(ndm1),DopCase)−1)×ΔFD) and coherent addition value Pow_ADDALIAS(f_(b_cfar),f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) with considerationgiven to Doppler aliasing are calculated.

As described above, time-Doppler demultiplexer 404 performs coherentaddition processing on the outputs of Doppler analyzers 402, withconsideration given to the presence or absence of Doppler aliasing.

Time-Doppler demultiplexer 404 detects coherent addition values thatprovide maximum power for DopCase=1, . . . , N_(DM), based on coherentaddition value Pow_ADDCOH(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) and coherentaddition value Pow_ADDALIAS(f_(b_cfar),f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase)−1)×ΔFD) with considerationgiven to Doppler aliasing. In the following, the index number of DopCasethat provides a coherent addition value that realizes maximum power isrepresented by “DopCase_max”.

For example, when coherent addition value Pow_ADDCOH(f_(b_cfar),f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase_max)−1)×ΔFD) indicatesmaximum power, time-Doppler demultiplexer 404 determines “absence ofaliasing”.

For example, when coherent addition value Pow_ADDALIAS(f_(b_cfar),f_(s_comp_cfar)±(DOPposi(DOP_(ndm1), DopCase_max)−1)×ΔFD) withconsideration given to Doppler aliasing indicates maximum power,time-Doppler demultiplexer 404 determines “presence of aliasing”.

A reduction in the SNR of a reception signal that is a reflected wavemay make it difficult to discriminate detection of the maximum coherentaddition power level due to the influence of noise. To address thisdifficulty, time-Doppler demultiplexer 404 may introduce a determinationcondition and perform a process of, for example, adopting adetermination result (in other words, detection result) when thedetermination condition is satisfied and removing (in other words, notadopting) a determination result when the determination condition is notsatisfied. This can reduce the probability of erroneous detection of thenoise component and the like. For example, time-Doppler demultiplexer404 may adopt, as a determination result, detection value P_(MAX) ofreceived power having maximum coherent addition value indicating thepresence or absence of Doppler aliasing when satisfyingP_(MAX)>LEV_(DETECT)×PowerFT(f_(b_cfar), f_(s_comp_cfar)) LEV_(DETECT)is a determination threshold. LEV_(DETECT) is a real number satisfying0<LEV_(DE0TECT)<1.

<(2) Process for Discrimination of Transmit Antennas 306 andDiscrimination of Doppler Frequencies of Target>

For example, time-Doppler demultiplexer 404 detects Doppler frequencyf_(TARGET) of the target in the range of −1/(2×Tr)≤f_(TARGET)<1/(2×Tr).

For example, time-Doppler demultiplexer 404 discriminates transmitantennas 306 and Doppler frequencies of the target, based on thedetermination result of DopCase (for example, DopCase_max) and thedetermination result of the presence or absence of aliasing.

[Case (a): Case without Aliasing]

Doppler Frequency Determination for Target:

Time-Doppler demultiplexer 404 determines that the Doppler frequencyindex of the target is (f_(s_comp_cfar)+(DopCase_max−1)×ΔFD−DOP₁). Forexample, the Doppler frequency interval for Doppler frequency indexf_(s_comp_cfar) is given by 1/(Ntdslot×N_(TD)×Tr). Therefore,time-Doppler demultiplexer 404 determines that Doppler frequencyf_(TARGET) of the target is (f_(s_comp_cfar)(DopCase_max−1)×ΔFD−DOP₁)/(Ntdslot×N_(TD)×Tr).

Transmit Antenna Determination:

Time-Doppler demultiplexer 404 determines, for transmit antenna 306 withthe number of time-Doppler multiplexes being 1, that, for example,coherent addition signal ADDCOH_(z)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase_max)−1)×ΔFD) for distanceindex f_(b_cfar) and Doppler frequency index(f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase_max)−1)×ΔFD) in the z-thsignal processor 206 is a reception signal for transmit antenna Tx #[1,ndm1].

Further, time-Doppler demultiplexer 404 determines, for transmit antenna306 with the number of time-Doppler multiplexes being N_(TD), that theoutput of Doppler analyzer 402 for distance index f_(b_cfar) and Dopplerfrequency index (f_(s_comp_cfar)+(DOPposi(DOP_(ndm),DopCase_max)−1)×ΔFD) (see, for example, following equations 117 and 118)in the z-th signal processor 206 is a reception signal for transmitantenna Tx #[ntd, ndm]. In equation 117, the exp term is a term forcorrecting the Doppler phase caused by time multiplexing. Further,ndm=1, . . . , N_(DM), but excluding ndm1. Further, ntd=1, . . . ,N_(TD).

$\begin{matrix}{{{VFT}_{z}^{ntd}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{ndm},{DopCase\_ max}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {{- j}\frac{2{\pi( f_{est\_ max} )}}{N_{tdslot}} \times \frac{{ntd} - 1}{N_{TD}}} \}} & ( {{Equation}117} )\end{matrix}$ $\begin{matrix}{f_{est\_ max} = {f_{{s\_ comp}{\_ cfar}} + {( {{DopCase\_ max} - 1} ) \times \Delta{FD}} - {DOP}_{1}}} & ( {{Equation}118} )\end{matrix}$

[Case (b): Case with Abasing]

Doppler Frequency Determination for Target:

Time-Doppler demultiplexer 404 determines that the Doppler frequencyindex of the target is f_(s_comp_cfar)(DopCase_max−1)×ΔFD)−DOP₁−Ncode×Sign(f_(s_comp_cfar)(DopCase_max−1)×ΔFD−DOP₁). For example, the Doppler frequency intervalfor Doppler frequency index f_(s_comp_cfar) is given by1/(Ntdslot×N_(TD)×Tr). Therefore, time-Doppler demultiplexer 404determines that Doppler frequency f_(TARGET) of the target is(f_(s_comp_cfar)+(DopCase_max−1)×ΔFD−DOP₁−Ntdslot×Sign(f_(s_comp_cfar)+(DopCase_max−1)×ΔFD−DOP₁))/(Ntdslot×N_(TD)×Tr).Sign(x) is a sign function and is a function for real numbers x,providing as output 1 when x>0, 0 when x=0, and −1 when x<0.

In this manner, when the outputs of Doppler analyzers 402 containDoppler aliasing, time-Doppler demultiplexer 404 determines a Dopplerfrequency of the target, with consideration given to the aliasingcomponent (for example,Ntdslot×Sign(f_(s_comp_cfar)+(DopCase_max−1)×ΔFD−DOP₁)).

Transmit Antenna Determination:

Time-Doppler demultiplexer 404 determines, for transmit antenna 306 withthe number of time-Doppler multiplexes being 1, that, for example,coherent addition signal ADDALIAS_(z)(f_(b_cfar),f_(s_comp_cfar)+(DOPposi(DOP_(ndm1), DopCase_max)−1)×ΔFD) withconsideration given to aliasing for distance index f_(b_cfar) andDoppler frequency index (f_(s_comp_cfar)+(DOPposi(DOP_(ndm1),DopCase_max)−1)×ΔFD) in the z-th signal processor 206 is a receptionsignal for transmit antenna Tx #[1, ndm1].

Further, time-Doppler demultiplexer 404 determines, for transmit antenna306 with the number of time-Doppler multiplexes being N_(TD), that, forexample, the output of Doppler analyzer 402 for distance indexf_(b_cfar) and Doppler frequency index(f_(s_comp_cfar)±(DOPposi(DOP_(ndm), DopCase_max)−1)×ΔFD) (see, forexample, following equations 119 and 120) in the z-th signal processor206 is a reception signal for transmit antenna Tx #[ntd, ndm]. Inequation 119, the exp term is a term for correcting the Doppler phasecaused by time multiplexing. Further, ndm=1, . . . , N_(DM), butexcluding ndm1. Further, ntd=1, . . . , N_(TD).

$\begin{matrix}{{{VFT}_{z}^{ntd}( {f_{b\_ cfar},{f_{{s\_ comp}{\_ cfar}} + {\{ {{{DOPposi}( {{DOP}_{ndm},{DopCase\_ max}} )} - 1} \} \times \Delta{FD}}}} )} \times \exp\{ {j\frac{2{\pi( f_{est\_ max} )}}{N_{tdslot}} \times \frac{{ntd} - 1}{N_{TD}}} \} \times \text{ }\exp\{ {{- j}\frac{2\pi{{Sign}( f_{est\_ max} )}( {{ntd} - 1} )}{N_{TD}}} \}} & ( {{Equation}119} )\end{matrix}$ $\begin{matrix}{f_{est\_ max} = {f_{{s\_ comp}{\_ cfar}} + {( {{DopCase\_ max} - 1} ) \times \Delta{FD}} - {DOP}_{1}}} & ( {{Equation}120} )\end{matrix}$

As described above, radar apparatus 20 does not set all of N_(DM)numbers of time-Doppler multiplexes N_(DOP_TD)(1), N_(DOP_TD)(2), . . ., and N_(DOP_TD)(N_(DM)) to N_(TD), but sets at least one of the numbersof time-Doppler multiplexes to 1 (in other words, no time multiplexing).With this setting, for example, a time-Doppler multiplexed signal forwhich the number of time-Doppler multiplexes is set to N_(TD) istransmitted at a set transmission timing (for example, timing oftransmission on), whereas a time-Doppler multiplexed signal for whichthe number of time-Doppler multiplexes is set to 1 is transmitted foreach transmission period, for example. Therefore, a reception signal ofa time-Doppler multiplexed signal for which the number of time-Dopplermultiplexes is set to 1 has a maximum value for a coherent additionvalue with consideration given to the presence or absence of Doppleraliasing. Time-Doppler demultiplexer 404 can utilize this characteristicto discriminate transmit antennas 306 and to determine a Dopplerfrequency of the target including the presence or absence of aliasing.

Accordingly, in this embodiment, the range over which a Dopplerfrequency is detectable without ambiguity can be extended to the rangegreater than or equal to −1/(2Tr) and less than 1/(2Tr). For example,when a single antenna is used for transmission, the range over which aDoppler frequency is detectable without ambiguity is the range greaterthan or equal to −1/(2Tr) and less than 1/(2Tr). In this embodiment,therefore, even when plural transmit antennas 306 are used, radarapparatus 20 can detect a Doppler frequency without ambiguity in amanner similar to that when a single antenna is used for transmission.

The foregoing description has been given of an example of the operationof time-Doppler demultiplexer 404.

In FIG. 24 , direction estimator 405 performs a direction estimationprocess for the target based on the Doppler frequency determinationresult for the target input from time-Doppler demultiplexer 404, thetransmit antenna determination result (determination result of Doppleraliasing), the output of Doppler analyzer 402 for distance indexf_(b_cfar) and Doppler frequency index(f_(s_comp_cfar)+(DOPposi(DOP_(ndm), DopCase_max)−1)×ΔFD), and acoherent addition value that is a maximum value.

For example, direction estimator 405 generates virtual receive arraycorrelation vector h(f_(b_cfar), f_(s_comp_cfar)), based on the Dopplerfrequency determination result for the target and the transmit antennadetermination result, and performs a direction estimation process.

As described above, in this embodiment, radar apparatus 20 applies phaserotation amounts corresponding to Doppler shift amounts to radartransmission signals and time-multiplexes radar transmission signals(for example, Doppler multiplexed signals) to transmit radartransmission signals (for example, time-Doppler multiplexed signals)from plural transmit antennas 108 in a multiplexed manner. In thisembodiment, plural transmit antennas 108 are associated withcombinations of Doppler shift amounts (DOP_(ndm)) and transmissiondurations for time multiplexing (for example, transmission timings) suchthat in each of the combinations, at least one of Doppler shift amount(DOP_(ndm)) or the transmission duration for time multiplexing (forexample, transmission timing) is different. In this embodiment,furthermore, the number of time multiplexes corresponding to eachDoppler shift amount in combinations of Doppler shift amounts andtransmission durations for time multiplexing is different. In otherwords, the numbers of time multiplexes for the respectiveDoppler-multiplexed transmission signals are set to be non-uniform.

Radar apparatus 20 can determine, based on, for example, a receptionsignal of each time-Doppler multiplexed signal (for example, coherentaddition signal), transmit antenna 306 associated with the time-Dopplermultiplexed signal (in other words, the combination of Doppler shiftamount and transmission duration) and the presence or absence of Doppleraliasing (for example, DopCase and the like). This enables radarapparatus 20 to appropriately determine a Doppler frequency of thetarget even in the presence of Doppler aliasing.

According to this embodiment, therefore, radar apparatus 20 can extendthe effective Doppler frequency bandwidth to 1/(Tr) and can extend theDoppler frequency (relative velocity) detection range with no ambiguity.Accordingly, radar apparatus 20 can improve target-object sensingaccuracy over a wider Doppler frequency range.

In this embodiment, furthermore, time-Doppler multiplexing, which isperformed using both Doppler multiplexing and time multiplexing, canreduce the number of Doppler multiplexes compared with the use of onlyDoppler multiplexing in multiplexing transmission. This can increase theintervals of phase rotation amounts for applying Doppler shifts,thereby, for example, relieving the accuracy requirements (phasemodulation accuracy) for the phase shifters and achieving the costreduction effect of an RF section, including reduction of the man-hoursrequired for the adjustment of the phase shifters.

In this embodiment, furthermore, since time-Doppler multiplexing isperformed using both Doppler multiplexing and time multiplexing, radarapparatus 20 performs, for each time multiplexing index, Fourierfrequency analysis (FFT processing) to detect the Doppler frequency(detect the relative velocity). Accordingly, for example, compared withFourier frequency analysis (FFT processing) to detect the Dopplerfrequency (detect the relative velocity) using only Doppler multiplexingin multiplexing transmission, the FFT size is (1/number of timemultiplexes N_(TD)) and the number of FFTs is increased by (the numberof time multiplexes N_(TD)) times. According to this embodiment,therefore, the operation reduction effect of FFT processing can beachieved, and the effect of simplification of the circuit configurationand cost reduction can also be achieved.

The foregoing description has been given of an exemplary embodiment ofthe present disclosure.

OTHER EMBODIMENTS

For example, Embodiment 1 has described the case where in an example inwhich the number of code multiplexes is set to be smaller than N_(CM),the number of coded Doppler multiplexes is 1 (in other words, no codemultiplexing). However, in an exemplary embodiment of the presentdisclosure, for example, when the number of coded Doppler multiplexes isset to be smaller than N_(CM), the number of coded Doppler multiplexesmay be set to be greater than or equal to 2.

In a radar apparatus according to an exemplary embodiment of the presentdisclosure, a radar transmitter and a radar receiver may be individuallyarranged in physically separate locations. In a radar receiver accordingto an exemplary embodiment of the present disclosure, a directionestimator and any other component may be individually arranged inphysically separate locations.

Further, in an exemplary embodiment of the present disclosure, thevalues used (such as the number Nt of transmit antennas, the number Naof receive antennas, the number of Doppler multiplexes N_(DM), thenumber of codes N_(CM), and the number of time multiplexes N_(TD)) areexamples, and these values are not restrictive.

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes, for example, a central processing unit (CPU), astorage medium such as a read only memory (ROM) that stores a controlprogram, and a work memory such as a random access memory (RAM), whichare not illustrated. In this case, the functions of the sectionsdescribed above are implemented by the CPU executing the controlprogram. However, the hardware configuration of the radar apparatus isnot limited to that in this example. For example, the functionalsections of the radar apparatus may be implemented as an integratedcircuit (IC). Each functional section may be formed as an individualchip, or some or all of them may be formed into a single chip. In theforegoing description, the term “section” used to indicate eachconstituent element may be interchangeably referred to as any other termsuch as “circuit (circuitry)”, “device”, “unit”, or “module”.

Various embodiments have been described with reference to the drawingshereinabove. Obviously, the present disclosure is not limited to theseexamples. Obviously, a person skilled in the art would arrive variationsand modification examples within a scope described in claims, and it isunderstood that these variations and modifications are within thetechnical scope of the present disclosure. Each constituent element ofthe above-mentioned embodiments may be combined optionally withoutdeparting from the spirit of the disclosure.

The above embodiments have been described with an example of aconfiguration using hardware, but the present disclosure can be realizedby software in cooperation with hardware.

Each functional block used in the description of each embodimentdescribed above is typically realized by an LSI, which is an integratedcircuit. The integrated circuit controls each functional block used inthe description of the above embodiments and may include an inputterminal and an output terminal. The LSI may be individually formed aschips, or one chip may be formed so as to include a part or all of thefunctional blocks. The LSI herein may be referred to as an IC, a systemLSI, a super LSI, or an ultra LSI depending on a difference in thedegree of integration.

However, the technique of implementing an integrated circuit is notlimited to the LSI and may be realized by using a dedicated circuit, ageneral-purpose processor, or a special-purpose processor. In addition,a Field Programmable Gate Array (FPGA) that can be programmed after themanufacture of the LSI or a reconfigurable processor in which theconnections and the settings of circuit cells disposed inside the LSIcan be reconfigured may be used.

If future integrated circuit technology replaces LSIs as a result of theadvancement of semiconductor technology or other derivative technology,the functional blocks could be integrated using the future integratedcircuit technology. Biotechnology can also be applied.

SUMMARY OF THE DISCLOSURE

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes a plurality of transmit antennas that transmit aplurality of transmission signals using a multiplexing transmission, anda transmission circuit that applies phase rotation amounts correspondingto combinations of Doppler shift amounts and code sequences to theplurality of transmission signals, wherein each of the combinations ofthe Doppler shift amounts and the code sequences has at least onedifferent from other combination, and wherein the number of multiplexesof the code sequence corresponding to at least one of the Doppler shiftamounts in the combinations is different from the number of multiplexingof code sequences corresponding to the remaining Doppler shift amount.

In an exemplary embodiment of the present disclosure, the number ofphases used for phase rotation amounts for applying the Doppler shiftamounts is smaller than the number of the plurality of transmitantennas.

In an exemplary embodiment of the present disclosure, intervals of phaserotation amounts for applying the Doppler shift amounts are equal.

In an exemplary embodiment of the present disclosure, the number ofphases used for phase rotation amounts for applying the Doppler shiftamounts is equal to the number of the Doppler shift amounts that areused for the multiplexing transmission.

In an exemplary embodiment of the present disclosure, the radarapparatus further includes a plurality of receive antennas that receivereflected wave signals, the reflected wave signals being the pluralityof transmission signals reflected from a target, and reception circuitthat determines a transmit antenna corresponding to each of thereflected wave signals among the plurality of transmit antennas anddetermines whether the reflected wave signal contains aliasing in aDoppler frequency domain, based on a signal subjected to code separationfor the reflected wave signal using a corresponding one of the codesequences.

In an exemplary embodiment of the present disclosure, the transmitantennas have a sub-array configuration.

In an exemplary embodiment of the present disclosure, the transmissioncircuit sets intervals of the Doppler shift amounts to vary for each offrames in which the plurality of transmission signals are transmitted.

In an exemplary embodiment of the present disclosure, the transmissioncircuit sets transmission periods of the plurality of transmissionsignals to vary for each of frames in which the plurality oftransmission signals are transmitted.

In an exemplary embodiment of the present disclosure, the transmissioncircuit multiplies each of the plurality of transmission signals by apseudo-random code sequence.

In an exemplary embodiment of the present disclosure, the transmissioncircuit sets an association between each of the plurality of transmitantennas and one of the combinations of the Doppler shift amounts andthe code sequences to vary for each of frames in which the plurality oftransmission signals are transmitted.

In an exemplary embodiment of the present disclosure, the transmissioncircuit sets an association between each of the plurality of transmitantennas and one of the Doppler shift amounts to vary for each oftransmission periods of the plurality of transmission signals.

In an exemplary embodiment of the present disclosure, a radar apparatusincludes a plurality of transmit antennas that transmit transmissionsignals, and a transmission circuit that applies phase rotation amountsto the transmission signals to perform multiplexing transmission totransmit the transmission signals from the plurality of transmitantennas in a multiplexed manner, wherein the number of phases used forthe phase rotation amounts is smaller than the number of the pluralityof transmit antennas.

In an exemplary embodiment of the present disclosure, the number of theplurality of transmit antennas is any of 3, 6, and 7.

In an exemplary embodiment of the present disclosure, a radar apparatusincludes a plurality of transmit antennas that transmit transmissionsignals, and a transmission circuit that applies phase rotation amountscorresponding to Doppler shift amounts to the transmission signals andtime-multiplexes the transmission signals to perform multiplexingtransmission to transmit the transmission signals from the plurality oftransmit antennas in a multiplexed manner, wherein the plurality oftransmit antennas are associated with combinations of the Doppler shiftamounts and transmission durations for the time multiplexing such thatin each of the combinations, at least one of the Doppler shift amount orthe transmission duration for the time multiplexing is different, andthe number of multiplexes for the time multiplexing corresponding toeach of the Doppler shift amounts in the combinations is different.

In an exemplary embodiment of the present disclosure, the number ofphases used for phase rotation amounts for applying the Doppler shiftamounts is equal to the number of the Doppler shift amounts that areused for the multiplexing transmission.

In an exemplary embodiment of the present disclosure, intervals of phaserotation amounts for applying the Doppler shift amounts are equal.

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes a plurality of transmit antennas that transmittransmission signals, and a transmission circuit that applies phaserotation amounts corresponding to Doppler shift amounts and codesequences to the transmission signals to perform multiplexingtransmission to transmit the transmission signals from the plurality oftransmit antennas in a multiplexed manner, wherein each of thetransmission signals is a chirp signal, and a center frequency of thechirp signal is changed for each of transmission periods of thetransmission signals or each of transmission periods of the codesequences.

A radar apparatus according to an exemplary embodiment of the presentdisclosure includes a plurality of transmit antennas that transmittransmission signals, and a transmission circuit that applies phaserotation amounts corresponding to Doppler shift amounts and codesequences to the transmission signals to perform multiplexingtransmission to transmit the transmission signals from the plurality oftransmit antennas in a multiplexed manner, wherein the transmissionsignals are chirp signals, and center frequencies of the chirp signalsare changed in a round in a period that is a divisor multiple of a codelength of each of the code sequences relative to each of transmissionperiods of the transmission signals.

INDUSTRIAL APPLICABILITY

While various embodiments have been described herein above, it is to beappreciated that various changes in form and detail may be made withoutdeparting from the sprit and scope of the invention(s) presently orhereafter claimed.

This application is entitled and claims the benefit of Japanese PatentApplication No. 2019-110522, filed on Jun. 13, 2019 and Japanese PatentApplication No. 2019-221249, filed on Dec. 6, 2019, the disclosure ofwhich including the specification, drawings and abstract is incorporatedherein by reference in its entirety.

The present disclosure is suitable as a radar apparatus for wide-anglerange sensing.

REFERENCE SIGNS LIST

-   10, 10 b, 10 c, 10 d, 20 Radar apparatus-   100, 100 a, 100 b, 100 c, 100 d, 300 Radar transmitter-   101, 101 c, 101 d Radar transmission signal generator-   102 Modulated signal emitter-   103, 103 c, 103 d VCO-   104, 104 b, 104 d, 301 Phase rotation amount setter-   105, 302 Doppler shift setter-   106, 106 d Coder-   107, 304 Phase rotator-   108, 306 Transmit antenna-   109 Beam weight generator-   110 Beam weight multiplier-   111 Random code applier-   112, 112 d Transmission frequency controller-   200, 200 b, 200 c, 200 d, 400 Radar receiver-   201 Antenna channel processor-   202 Receive antenna-   203 Receiving radio section-   204 Mixer section-   205 LPF-   206, 206 b Signal processor-   207 AD converter-   208 Beat frequency analyzer-   209, 401 Output switching section-   210, 402 Doppler analyzer-   211, 403 CFAR section-   212, 212 d Coded Doppler demultiplexer-   213, 213 c, 213 d, 405 Direction estimator-   214 Random code multiplier-   303 Time multiplexer-   305 Transmission controller-   404 Time-Doppler demultiplexer

What is claimed is:
 1. A radar apparatus, comprising: a plurality oftransmit antennas that transmits a plurality of transmission signalsusing a multiplexing transmission; and a transmission circuit thatapplies phase rotation amounts corresponding to combinations of Dopplershift amounts and code sequences to the plurality of transmissionsignals, wherein each of the plurality of transmission signals isassigned a different combination among the combinations, and wherein thecombinations include at least one combination of different numbers ofmultiplexing by the code sequences.
 2. The radar apparatus according toclaim 1, wherein a number of phases used for the phase rotation amountsfor applying the Doppler shift amounts is smaller than a number of theplurality of transmit antennas.
 3. The radar apparatus according toclaim 1, wherein intervals of phase rotation amounts for applying theDoppler shift amounts are equal to each other.
 4. The radar apparatusaccording to claim 1, wherein a number of phases used for the phaserotation amounts for applying the Doppler shift amounts is equal to anumber of the Doppler shift amounts that are used for the multiplexingtransmission.
 5. The radar apparatus according to claim 1, furthercomprising: a plurality of receive antennas that receives reflected wavesignals, the reflected wave signals being the plurality of transmissionsignals reflected from a target; and a reception circuit that determinesa transmit antenna corresponding to each of the reflected wave signalsamong the plurality of transmit antennas and determines whether thereflected wave signal contains aliasing in a Doppler frequency domain,based on a signal subjected to code separation for the reflected wavesignal using a corresponding one of the code sequences.
 6. The radarapparatus according to claim 1, wherein the plurality of transmitantennas has a sub-array configuration.
 7. The radar apparatus accordingto claim 1, wherein the transmission circuit sets intervals of theDoppler shift amounts to vary for each of frames in which the pluralityof transmission signals is transmitted.
 8. The radar apparatus accordingto claim 1, wherein the transmission circuit sets transmission periodsof the plurality of transmission signals to vary for each of frames inwhich the plurality of transmission signals is transmitted.
 9. The radarapparatus according to claim 1, wherein the transmission circuitmultiplies each of the plurality of transmission signals by apseudo-random code sequence.
 10. The radar apparatus according to claim1, wherein the transmission circuit sets an association between each ofthe plurality of transmit antennas and one of the combinations of theDoppler shift amounts and the code sequences to vary for each of framesin which the plurality of transmission signals is transmitted.
 11. Theradar apparatus according to claim 1, wherein the transmission circuitsets an association between each of the plurality of transmit antennasand one of the Doppler shift amounts to vary for each of transmissionperiods of the plurality of transmission signals.
 12. A radar apparatus,comprising: a plurality of transmit antennas that transmits a pluralityof transmission signals using a multiplexing transmission; and atransmission circuit that applies phase rotation amounts correspondingto combinations of Doppler shift amounts and code sequences to theplurality of transmission signals.